Reception apparatus, phase error estimation method, and phase error correction method

ABSTRACT

In a phase error corrector, a signal extractor extracts received reference signals from received signals, and an error vector calculator calculates the error vectors of phase errors by comparing the extracted received reference signals with a known reference signal that is to be transmitted. A representative vector calculator divides, according to frequency, the error vectors into two or more groups and calculates representative vectors for the respective groups. A correction value calculator calculates, on the basis of the representative vectors, phase correction values for the respective frequencies. A phase corrector uses the calculated phase correction values to correct the phase errors for the respective frequencies.

TECHNICAL FIELD

The present disclosure relates to a reception apparatus, a phase errorestimation method, and a phase error correction method in wirelesscommunication.

BACKGROUND ART

In wireless communication systems, a carrier frequency error and asymbol synchronization shift caused between a transmitter and a receiverare corrected. For example, in a wireless communication apparatuscompliant with an OFDM (Orthogonal Frequency Division Multiplexing)wireless communication method, as a carrier frequency error correctionmethod, normally, a method is adopted where rough frequency correctionand rough symbol synchronization shift correction are performed andthen, a phase error caused by a residual carrier frequency offset and aresidual symbol synchronization shift is estimated and corrected.

Examples of conventional phase error estimation methods include a methodshown in Patent Document 1. Moreover, examples of conventional phaseerror correction methods include methods shown in Patent Documents 2 and3.

PRIOR ART DOCUMENTS Patent Documents

Patent Document 1: JP-A-2004-104744

Patent Document 2: JP-T-2012-519986

Patent Document 3: WO2008/047776

SUMMARY OF THE INVENTION Problem to be Solved by the Invention

An object of the present disclosure is to provide a reception apparatusand a phase error estimation method capable of obtaining a highlyprecise phase correction value even when the phase error or the noiselevel is high.

Another object of the present disclosure is to provide a receptionapparatus and a phase error estimation method capable of appropriatelycorrecting the carrier frequency offset and making it possible tocorrectly demodulate the received data.

Means for Solving the Problem

The present disclosure provides a reception apparatus having: a phaseerror estimator that extracts a specific reference signal from areceived signal which is a received transmitted signal having thespecific reference signal and obtains a received reference signal in afrequency domain in a receiver, compares, for each frequency, thereceived reference signal in the frequency domain and a transmittedreference signal which is the specific reference signal in thetransmitted signal expressed in the frequency domain and obtains aplurality of error vectors, and obtains, from the error vectors, a phaseerror inclination and a phase error offset in the frequency domainpossessed by the received reference signal and estimates a phase erroraccording to the frequency by the phase error inclination and the phaseerror offset; and a phase error corrector that corrects the phase errorfor the received signal by using a phase error estimation value obtainedby the phase error estimator.

The present disclosure provides a phase error estimation method of areception apparatus, having: extracting a specific reference signal froma received signal which is a received transmitted signal having thespecific reference signal and obtaining a received reference signal in afrequency domain in a receiver; comparing, for each frequency, thereceived reference signal in the frequency domain and a transmittedreference signal which is the specific reference signal in thetransmitted signal expressed in the frequency domain and obtaining aplurality of error vectors; dividing the error vectors into not lessthan two groups, obtaining a representative value of each group andobtaining a plurality of representative vectors; and obtaining the phaseerror inclination and the phase error offset in the frequency domainpossessed by the received reference signal based on the representativevectors and estimating a phase error according to the frequency by thephase error inclination and the phase error offset.

The present disclosure provides a phase error correction method of areception apparatus, having: extracting a specific reference signal froma received signal which is a received transmitted signal having thespecific reference signal and obtaining a received reference signal in afrequency domain after rough carrier frequency offset correction andrough symbol synchronization shift correction in a receiver; comparing,for each frequency, the received reference signal in the frequencydomain and a transmitted reference signal which is the specificreference signal in the transmitted signal expressed in the frequencydomain and obtaining a plurality of error vectors; estimating a phaseerror by linear approximation in the frequency domain from the errorvectors and obtaining the phase error inclination as a residual symbolsynchronization shift and the phase error offset as a residual carrierfrequency offset; performing correction of the residual symbolsynchronization shift in the frequency domain; and performing correctionof the residual carrier frequency offset on the received signalconverted from the frequency domain into the time domain.

The present disclosure provides a phase error correction method of areception apparatus, having: extracting a specific reference signal froma received signal which is a received transmitted signal having thespecific reference signal and obtaining a received reference signal in afrequency domain after rough carrier frequency offset correction andrough symbol synchronization shift correction in a receiver; comparing,for each frequency, the received reference signal in the frequencydomain and a transmitted reference signal which is the specificreference signal in the transmitted signal expressed in the frequencydomain and obtaining a plurality of error vectors; estimating a phaseerror by linear approximation in the frequency domain from the errorvectors and obtaining the phase error inclination as a residual symbolsynchronization shift and the phase error offset as a residual carrierfrequency offset; and performing correction of the residual symbolsynchronization shift and correction of the residual carrier frequencyoffset on the received signal converted from the frequency domain intothe time domain.

Advantage of the Invention

According to the present disclosure, a highly precise phase correctionvalue can be obtained even when the phase error or the noise level ishigh.

According to the present disclosure, the carrier frequency offset can beappropriately corrected and it can be made possible to correctlydemodulate the received data.

BRIEF DESCRIPTION OF THE DRAWINGS

[FIG. 1] A block diagram showing the structure of a receiver of awireless communication apparatus compliant with the WiGig.

[FIG. 2] A view showing a signal format of the WiGig.

[FIG. 3] A view showing a residual carrier frequency offset and aresidual symbol synchronization shift in the frequency domain.

[FIG. 4] A view showing an example of the occurrence of phasediscontinuity in the frequency domain.

[FIG. 5] A block diagram showing the structure of a receiver of awireless communication apparatus using the conventional phase errorestimation method.

[FIG. 6] A block diagram showing the structure of a phase errorcorrector in a first embodiment.

[FIG. 7] A view showing the spectrum of a reference signal GI in thefrequency domain and the frequency number used for the error vectorcalculation.

[FIG. 8] A view showing the structure of an error vector calculator.

[FIG. 9] A view showing the structure of a representative vectorcalculator.

[FIG. 10] A view showing the phase errors due to the residual carrierfrequency offset and the residual symbol synchronization shift in thefrequency domain in the case of a low SNR.

[FIG. 11] A view showing the error vectors of the phase errors in thecase of the low SNR.

[FIG. 12] A view showing the representative vectors of the error vectorsof the phase errors.

[FIG. 13] A view showing the phase errors of the low frequencyrepresentative vector and the high frequency representative vector.

[FIG. 14] A view showing the structure of a correction value calculator.

[FIG. 15] A view showing the relationship between normaladdition/subtraction and wrapping addition/subtraction.

[FIG. 16] A view showing the correspondence between the phases and thenumerical values when the phases of ±π are assigned to four-bit values.

[FIG. 17] A view showing the structure of a determiner of a phase offsetcalculator in the correction value calculator.

[FIG. 18] A view showing the structure of a frequency-by-frequencycorrection value calculator in the correction value calculator.

[FIG. 19] A view showing the structure of a phase corrector.

[FIG. 20] A view showing an example of the result of the phase errorestimation by a phase error corrector of the present embodiment.

[FIG. 21] A block diagram showing the structure where the phase errorcorrector of the present embodiment is applied to a receiver of awireless communication apparatus compliant with the OFDM.

[FIG. 22] A view showing the spectrum of the reference signal GI in thefrequency domain and the frequency number specifying the low frequencydomain and the high frequency domain according to a second embodiment.

[FIG. 23] A block diagram showing the structure of a phase errorcorrector in a third embodiment.

[FIG. 24] A view showing the phase errors due to the residual carrierfrequency offset and the residual symbol synchronization shift in thefrequency domain in the case of a low SNR.

[FIG. 25] A view showing the error vectors of the phase errors in thecase of the low SNR.

[FIG. 26] A view showing the representative vectors of the error vectorsof the phase errors.

[FIG. 27] A view showing the structure of a correction value calculatorin the third embodiment.

[FIG. 28] A block diagram showing the structure of a correction valuecalculator in a fourth embodiment.

[FIG. 29] A block diagram showing the structure of a receiver of awireless communication apparatus in a fifth embodiment.

[FIG. 30] A block diagram showing the structure of a phase errorcorrector in the fifth embodiment.

[FIG. 31] A view showing the structure of a transmission path correctorin the fifth embodiment.

[FIG. 32] A block diagram showing the structure of a receiver of awireless communication apparatus compliant with the WiGig.

[FIG. 33] A view showing a signal format of the WiGig.

[FIG. 34] A view showing the Ga mutual correlation peak in the STF.

[FIG. 35] A block diagram showing the structure of a receiver of awireless communication apparatus using the conventional residual carrierfrequency offset correction method.

[FIG. 36] A view showing the carrier frequency offset correction timingin the conventional example.

[FIG. 37] A block diagram showing the structure of a receiver of awireless communication apparatus in a sixth embodiment.

[FIG. 38] A block diagram showing the structure of a phase errorcorrector in the sixth embodiment.

[FIG. 39] A view showing the structure of an error vector calculator.

[FIG. 40] A view showing the structure of a phase error calculator.

[FIG. 41] A view showing the structure of a residual symbolsynchronization shift calculator.

[FIG. 42] A view showing the structure of a residual symbolsynchronization shift corrector.

[FIG. 43] A view showing the structure of a residual carrier frequencyoffset corrector.

[FIG. 44] A view showing the carrier frequency offset correction timingin the present embodiment.

[FIG. 45] A block diagram showing the structure of a residual carrierfrequency offset corrector in a seventh embodiment.

[FIG. 46] A block diagram showing the structure of a residual symbolsynchronization shift corrector in an eighth embodiment.

[FIG. 47] A block diagram showing the structure of a receiver of awireless communication apparatus in a ninth embodiment.

[FIG. 48] A block diagram showing the structure of a receiver of awireless communication apparatus in a tenth embodiment.

[FIG. 49] A view showing the structure of a phase error estimator.

[FIG. 50] A view showing the structure of a time domain residual symbolsynchronization shift corrector.

[FIG. 51] A view showing the relationship between the phase errorinclination and the residual symbol synchronization shift correctionvalue.

[FIG. 52] A view showing an example of the correction coefficientselection.

[FIG. 53] A block diagram showing the structure of a receiver of awireless communication apparatus in an eleventh embodiment.

[FIG. 54] A schematic view showing the rotation of the phase of themaximum correlation value by a carrier frequency error when the carrierfrequency error is present and a sampling frequency error is absentbetween the transmission and reception apparatuses.

[FIG. 55] A schematic view showing the rotation of the phase of themaximum correlation value by a carrier frequency error when the carrierfrequency error and a sampling frequency error are present between thetransmission and reception apparatuses.

[FIG. 56] A block diagram showing a structure example of a receptionapparatus in a twelfth embodiment.

[FIG. 57] A schematic view showing a first example of the structure of aphase rotator in the twelfth embodiment.

[FIG. 58] A schematic view showing a second example of the structure ofa phase rotator in the twelfth embodiment.

[FIG. 59] A schematic view showing a third example of the structure of aphase rotator in the twelfth embodiment.

[FIG. 60] A schematic view showing an example of the correlation valueof each sample in the twelfth embodiment.

[FIG. 61] (A) to (C) are schematic views showing an example ofcharacteristics of the maximum correction value in a case where no phaserotator is present and Fs_t=Fs_r.

[FIG. 62] (A) to (C) are schematic views showing an example ofcharacteristics of the maximum correction value in a case where no phaserotator is present and Fs_t<Fs_r.

[FIG. 63] (A) to (C) are schematic views showing an example ofcharacteristics of the maximum correction value in a case where no phaserotator is present and Fs_t>Fs_r.

[FIG. 64] A schematic view showing an example of the relationshipbetween the amount of phase rotation by the π/2 shift and the amount ofphase reverse rotation by the −π/2 shift in a case where Fs_t=Fs_r inthe twelfth embodiment.

[FIG. 65] A schematic view showing an example of the relationshipbetween the amount of phase rotation by the π/2 shift and the amount ofphase reverse rotation by the −π/2 shift in a case where Fs_t<Fs_r inthe twelfth embodiment.

[FIG. 66] A schematic view showing an example of the relationshipbetween the amount of phase rotation by the π/2 shift and the amount ofphase reverse rotation by the −π/2 shift in a case where Fs_t>Fs_r inthe twelfth embodiment.

[FIG. 67] A schematic view showing an example of correction of the phaseof the maximum correlation value in the twelfth embodiment.

[FIG. 68] A schematic view showing an example of characteristics of thephase change of the maximum correlation value and the phase change ofthe reference phase in the twelfth embodiment.

[FIG. 69] A schematic view showing a first example of the structure of afrequency corrector in the twelfth embodiment.

[FIG. 70] A schematic view showing a second example of the structure ofthe frequency corrector in the twelfth embodiment.

[FIG. 71] A block diagram showing a structure example of a receptionapparatus in a thirteenth embodiment.

[FIG. 72] A schematic view showing a structure example of a phaserotator in the thirteenth embodiment.

MODE FOR CARRYING OUT THE INVENTION Circumstances Leading up to theContents of an Embodiment of the Present Disclosure

In the phase error estimation and correction, if the phase error becomeslarge when a residual carrier frequency offset and a residual symbolsynchronization shift are corrected, there are cases where the phaseexceeds a range of −π to π [rad]. In this case, a discontinuous phasechange occurs as a phase error detected in the range of −π to π [rad].For this reason, the linearity of the phase with respect to thefrequency is spoiled, so that a phase correction value is not correctlyobtained.

For example, in a conventional example of the phase error estimationmethod shown in the above-mentioned Patent Document 1, when no phasediscontinuity is detected by the determiner, a correction value of theobtained phase error is calculated and phase correction is performed. Onthe other hand, when phase discontinuity is detected, phase correctionis performed by using the correction value obtained in the immediatelypreceding correction. Moreover, a method called phase unwrappingprocessing is generally known where when it is determined that theabsolute value of the difference between adjoining phases has exceededπ, the phases are made continuous by adding ±2π to one of the phases.

In the above-described conventional example, a problem arises in that acorrect correction value is not obtained in a situation where the phaseerror or the noise level is high. When the phase error or the noiselevel is high and phase discontinuity occurs, for example, in theearliest period, a correct correction value is not obtained by a methodof the conventional example using the immediately preceding correctionvalue. Moreover, even if a correct correction value is obtained in theearliest period, since the correction value is not updated if phasediscontinuity continuously occurs thereafter, the correction valuebecomes away from the one that has to be calculated. Moreover, when thephase unwrapping processing is used, there are cases where thedetermination as to whether the absolute value of the phase differencehas exceeded it or not is erroneously made because of the noisecontained in the received signal. In particular, in the wirelesscommunication standard WiGig (trademark; the same applies hereafter)(Wireless Gigabit) using a millimeter waveband, since the requirementfor the PER (Packet Error Rate) is strict, highly precise phase errorestimation is necessary. Accordingly, a phase error estimation methodfor obtaining a highly precise phase correction value even when thephase error or the noise level is high is desired.

The present disclosure shows an example of a wireless communicationapparatus including DFT (Discrete Fourier Transformation) and IDFT(Inverse Discrete Fourier Transformation) in a receiver such as OFDM(Orthogonal Frequency Division Multiplexing) like a wireless LANstandard IEEE 802.11a,g,n or SC-FDE (Single Carrier Frequency DomainEqualizer) like the WiGig.

Correction of a phase error caused between a transmitter and a receiverdue to a residual carrier frequency offset and a residual symbolsynchronization shift in a wireless communication apparatus of this kindwill be described below.

FIG. 1 is a block diagram showing the structure of a receiver of awireless communication apparatus compliant with the WiGig. The wirelesscommunication apparatus shown in FIG. 1 has an RF (Radio Frequency)processor 1, an ADC (Analog-Digital Converter) 2, a synchronizationdetector 4, a frequency corrector 5, an S/P (serial-parallel) converter6, a DFT section 7, a transmission path corrector 8, a phase errorcorrector 9, an IDFT section 10, a P/S (parallel-serial) converter 11, ademodulator 13 and a selector 15.

The RF processor 1 converts a received signal of a radio frequencyreceived by the antenna into a baseband signal which is a complexsignal. The ADC section 2 periodically samples the baseband signal whichis the complex signal, and converts it into a digital complex basebandsignal.

The synchronization detector 4 detects a known preamble signal forsynchronization (STF described later) from the complex baseband signal.The frequency corrector 5 calculates the error of the carrier frequencyby using the known preamble signal (STF described later), and performsrough carrier frequency offset correction. The S/P converter 6 convertsthe complex baseband signal which is a serial signal into a parallelsignal. The DFT section 7 converts the complex baseband signal in thetime domain having undergone the rough carrier frequency offsetcorrection, into a complex signal in the frequency domain after roughsymbol synchronization according to the timing of the preamble signaldetected by the synchronization detector 4.

The transmission path corrector 8 corrects the transmission path errorbetween the transmitter and the receiver by using a known preamblesignal (CEF described later). The selector 15 selects the output signalof the frequency corrector 5 or the P/S converter 11, and outputs it tothe transmission path corrector 8. The phase error corrector 9 corrects,by using a known reference signal (GI described later), a residual phaseerror caused by a residual carrier frequency offset and a residualsymbol synchronization shift.

The IDFT section 10 converts the phase-error-corrected signal in thefrequency domain outputted from the phase error corrector 9, into acomplex baseband signal in the time domain. The P/S converter 11converts the parallel signal which is the output of the IDFT section 10,into a serial signal. The demodulator 13 demodulates a digitallymodulated signal by using the complex baseband signal converted into thetime domain by the IDFT section 10.

FIG. 2 is a view showing a signal format of the WiGig. The signaltransmitted in a WiGig wireless communication system has, from the head,an STF (Short Training Field), a CEF (Channel Estimation Field), a GI(Guard Interval), a header (Header), . . . , and data portions (Data1,Data2, . . . ). Here, the STF and the CEF are provided as preamblesignals.

The STF is the repetition of the known preamble signal used in thesynchronization detector 4 and the frequency corrector 5 of FIG. 1.During the AGC period from the head of the STF, an AGC operation by anAGC section (not shown) is performed, and during the remaining rough CFOperiod, the calculation of the rough carrier frequency offset by thefrequency corrector 5 is performed. The last one symbol of the STF is asynchronization detection period when rough symbol synchronization isperformed by the detection of the preamble signal by the synchronizationdetector 4.

The CEF is the known preamble signal different from the above-describedSTF used in the transmission path corrector 8 of FIG. 1.

The header contains information representative of the attributes of thetransmission data such as the modulation method and the number oftransmitted symbols. The data portions contain the data to betransmitted itself. The GI is a known reference signal different fromthe above-mentioned STF and CEF and repetitively inserted at regulartime intervals in the headers and the data portions. The GI is used forthe phase error estimation (phase error correction value calculation) inthe phase error corrector 9 of FIG. 1.

Next, the residual carrier frequency offset and the residual symbolsynchronization shift corrected in the phase error corrector 9 will bedescribed. The carrier frequency offset is a phase error caused becausethe carrier frequency used when the complex baseband signal isorthogonally modulated in the RF processor of the transmitter (notshown) and the carrier frequency used for the orthogonal modulation inthe RF processor 1 of the receiver are subtly different.

The frequency corrector 5 estimates the error of the carrier frequency(rough carrier frequency offset) to perform correction, and since anerror is caused in the estimation of the carrier frequency offsetbecause of the influence of the signal noise and the phase noise of thecarrier, the phase error remains and accumulates. This is the residualcarrier frequency offset. For this reason, it is necessary to correctthe residual carrier frequency offset while continuously updating thecorrection value of the phase error.

The residual symbol synchronization shift is a phase error causedbecause the sampling frequency of the DAC (Digital Analog Converter)section generating the complex baseband signal in the transmitter (notshown) and the sampling frequency of the ADC section 2 of the receiverare subtly different. Because of the sampling frequency error betweenthe transmitter and the receiver, even if phase error correction isperformed in the earliest period, the symbol synchronization shiftremains and accumulates with the lapse of time, and the symbol timingerror increases. For this reason, it is necessary to correct theresidual symbol synchronization shift while continuously updating thephase error correction value.

In the DFT section 7, the timing of the DFT is controlled in accordancewith the head of the CEF by the preamble signal detected in thesynchronization detector 4. If the timing error of the symbol increases,the window synchronization in the DFT section 7 is shifted.

FIG. 3 is a view showing the residual carrier frequency offset and theresidual symbol synchronization shift in the frequency domain. In FIG.3, the horizontal axis shows the frequency, and the vertical axis showsthe phase error in the range of −π to π [rad]. As shown in the figure,since the phase error has linearity, the offset of the phase error (themean of the phase errors of the frequencies, that is, the offsetquantity at the mean value of the frequencies) represents the residualcarrier frequency offset, and the inclination of the phase error (theamount of phase error change with respect to the frequency) representsthe residual symbol synchronization shift.

Here, the phase error is the phase difference between the referencesignal GI extracted from the received signal and the known referencesignal GI to be transmitted. When the phase error is y, the frequency isx, the inclination of the phase error is a and the offset of the phaseerror is b, the phase error can be expressed by a linear interpolationformula like the following expression [1]:

y=a x+b   [1]

The phase error is obtained by performing linear approximation on themeasurement values of the phase errors at a plurality of frequencies.For the linear approximation, for example, the LSM (Least SquaresMethod) is used. The LSM processing can be realized by using a generallyknown method by approximation.

FIG. 4 is a view showing an example of the occurrence of phasediscontinuity in the frequency domain. When the phase error increases,as shown in the example of FIG. 4, the phase (8) changes from π [rad] to−π [rad], and a discontinuous phase change occurs. In this case, thelinearity of the phase with respect to the frequency is spoiled, so thatthe phase correction value cannot be obtained correctly.

Conventionally, as a measure to counter the discontinuous phase change,as shown in the above-described Patent Document 1, a method is adoptedwhere when phase discontinuity is detected, phase correction isperformed by using the correction value used in the immediatelypreceding correction. FIG. 5 is a block diagram showing the structure ofa receiver of a wireless communication apparatus using the conventionalphase error estimation method. In this conventional example, when nophase discontinuity is detected by a determiner 596, the phase errorcorrection value calculated in a correction value calculator 593 isoutputted from a correction value output section 595, and phasecorrection is performed in a phase corrector 594. On the other hand,when phase discontinuity is detected in the determiner 596, thecorrection value used in the immediately preceding correction isoutputted from the correction value output section 595 and phasecorrection is performed.

Moreover, a method is available where when it is determined that theabsolute value of the difference between adjoining phases has exceeded πby using the phase unwrapping processing, the phases are made continuousby adding ±2π to one of the phases. In the example of FIG. 4, unwrappingprocessing is performed where by a comparison between phases (7) and(8), the phase is returned by adding +2π to the phase (8).

In the conventional example, by the method using the immediatelypreceding correction value when phase discontinuity occurs, an erroneouscorrection value is used, for example, when phase discontinuity occursin the earliest period. Moreover, even if a correct correction value isobtained in the earliest period, since the correction value is notupdated if phase discontinuity continuously occurs thereafter, thecorrection value becomes away from the one that has to be calculated.Moreover, when the phase unwrapping processing is used, there are caseswhere the determination as to whether the absolute value of the phasedifference has exceeded it or not is erroneously made because of thenoise contained in the received signal.

These problems frequently arise when the phase error measurement valueis disturbed in a low SNR (Signal-to-Noise Ratio) situation. Forexample, in the wireless LAN standard, PER=10% is a requiredspecification, whereas in the WiGig, requirement for PER is as strict asPER=1% at a low SNR. In order that PER is not deteriorated, it isnecessary to realize more highly precise phase error correction.

In view of the above-mentioned problems, the present disclosure providesa reception apparatus and a phase error estimation method and apparatuscapable of easily and precisely determining phase discontinuity andobtaining a highly precise phase correction value when phasediscontinuity occurs in a phase error even when the phase error or thenoise level is high.

An Embodiment of the Present Disclosure

Hereinafter, an embodiment according to the present disclosure will bedescribed in detail with reference to the drawings. A receptionapparatus, a phase error estimation method and a phase error estimationapparatus according to the present disclosure are realized in a wirelesscommunication apparatus of the embodiment. With respect to the drawingsused in the following description, the same elements are denoted by thesame reference numerals and signs and overlapping descriptions areomitted.

First Embodiment

FIG. 6 is a block diagram showing the structure of a phase errorcorrector in a first embodiment of the present disclosure. Here, thestructure and the operation in the receiver of the wirelesscommunication apparatus compliant with the WiGig shown in FIG. 1 areillustrated.

In FIG. 1, the RF processor 1 amplifies a received signal of a radiofrequency received by the antenna, and performs orthogonal modulationthereon to convert it into a baseband signal. The baseband signal havingundergone the orthogonal modulation is a complex signal.

The ADC section 2 periodically samples the signal having undergone theorthogonal modulation in the RF processor 1, and converts it into adigital complex baseband signal.

The synchronization detector 4 detects the known preamble signal forsynchronization (STF) from the complex baseband signal, and outputs atiming signal for synchronization. The preamble signal is used forwindow synchronization, that is, rough symbol synchronization of the DFTsection 7.

The frequency corrector 5 calculates the rough carrier frequency offsetas the carrier frequency error by using the known preamble signal (STF),and outputs a complex baseband signal obtained by correcting the roughcarrier frequency offset.

The S/P converter 6 which is a buffer for operating the DFT section 7converts the complex baseband signal which is a serial signal into aparallel signal. The DFT section 7 which corresponds to an example of atime-frequency converter performs time-frequency conversion according tothe timing of the STF detected by the synchronization detector 4 withrespect to the complex baseband signal in the time domain havingundergone the rough carrier frequency offset correction, and outputs acomplex signal in the frequency domain.

The transmission path corrector 8 calculates the amplitude and the phasewhich are transmission characteristics possessed by the transmissionpath between the transmitter and the receiver, by using the knownpreamble signal (CEF), and corrects the transmission path error.

The selector 15 selects the signal of the frequency corrector 5 or theP/S converter 11, and outputs it to the transmission path corrector 8.The selection of the signal used for the transmission path errorcorrection is determined by the designer. By selecting the signal fromthe frequency corrector 5, the correction value can be calculatedfaster. By selecting the signal from the P/S converter 11, a correctionvalue considering the occurrence error between the DFT section 7 and theIDFT section 10 caused in circuitization can be calculated.

The phase error corrector 9 calculates the residual carrier frequencyoffset and the residual symbol synchronization shift by using theperiodically inserted known reference signal (GI) as a specificreference signal, and corrects the phase error due to the residualcarrier frequency offset and the residual symbol synchronization shiftin the frequency domain.

The IDFT section 10 which corresponds to an example of a frequency-timeconverter performs frequency-time conversion of the output signal of thephase error corrector 9, and converts it into a complex baseband signalin the time domain.

The P/S converter 11 converts the parallel signal which is the output ofthe IDFT section 10, into a serial signal.

The demodulator 13 demodulates a digitally modulated signal by using thecomplex baseband signal converted into the time domain and havingundergone the residual phase error correction, and obtains the receiveddata.

In the above-described structure, the synchronization detector 4, thefrequency corrector 5, the S/P converter 6, the DFT section 7, thetransmission path corrector 8, the phase error corrector 9, the IDFTsection 10, the P/S converter 11 and the demodulator 13 can beimplemented as an information processing circuit including a processorand a memory, and the functions can be realized by operating a softwareprogram in the processor to execute predetermined processing.

In FIG. 6, the phase error corrector 9 has a signal extractor 90, anerror vector calculator 91, a representative vector calculator 92, acorrection value calculator 93 and a phase corrector 94. Here, thesignal extractor 90, the error vector calculator 91, the representativevector calculator 92 and the correction value calculator 93 correspondto a phase error estimator 95.

In the signal extractor 90, in the frequency domain, the periodicallyand repetitively received reference signal (GI) (corresponding to anexample of the received reference signal) is extracted from the receivedsignal. In the error vector calculator 91, the reference signalextracted from the received signal and the known reference signal (GI)to be transmitted (corresponding to an example of the transmittedreference signal) are compared, and a plurality of error vectors due tothe difference therebetween are calculated.

In the representative vector calculator 92, the error vectors obtainedin the error vector calculator 91 are divided into not less than twogroups according to the frequency, and calculates the representativevalue of each group. Here, as an example of the representative value,the mean value of the error vectors of the groups is calculated andoutputted. As the mean value calculation method, a simple vector meanmay be used, or a predetermined weight may be assigned by the frequencyto calculate the mean. While a different value such as the median valueof each group may be used as the representative value, it is preferableto use the mean value in the situation assumed in the present embodimentwhere a phase error due to a random noise component is added under a lowSNR environment.

In the correction value calculator 93, the phase correction value ofeach frequency is calculated according to the frequency based on aplurality of representative vectors (vector mean values) of the groupsobtained in the representative vector calculator 92. In the phasecorrector 94, the phase error of each frequency is corrected by usingthe phase correction value calculated in the correction value calculator93.

FIG. 7 is a view showing the spectrum of the reference signal GI in thefrequency domain and the frequency number used for the error vectorcalculation. In FIG. 7, the horizontal axis shows the frequency numbercorresponding to each frequency, and the vertical axis shows theabsolute value of the amplitude of the spectrum of the GI.

In the signal extractor 90, the reference signal GI is extracted fromthe received signal, and the spectrum shown in FIG. 7 where thereference signal GI of 64 symbols is Fourier-transformed is obtained.Here, the frequency number is the number representative of eachfrequency where 27.5 MHz which is the quotient when 1.76 GHz (−880 MHzto +880 MHz) which is the symbol rate of the WiGig standard is dividedby 64 symbols is one unit.

Of the spectrum in the frequency domain, particularly, one with a highabsolute value is high in noise tolerance, and is hardly affected byphase noise. Therefore, here, as an example, a spectrum of apredetermined number (eight symbols in the illustrated example) indecreasing order of the absolute value of the amplitude is used as therepresentative value, and the frequency numbers −25, −22, −10 and −7,and 8, 13, 19 and 24 shown by black circles in FIG. 7 are furtherextracted.

FIG. 8 is a view showing the structure of the error vector calculator91. The error vector calculator 91 has complex multipliers 910-00 to910-07. Here, since the error vector of the reference signal GI iscalculated for eight frequencies, eight systems of circuits are providedin parallel. To calculate the error vector, the reference signalextracted from the received signal and the known reference signal to betransmitted are compared by using the complex multipliers 910-00 to910-07.

The complex multipliers 910-00 to 910-07 are supplied with coefficientsref00 to ref07 of the known reference signal serving as the reference,respectively, and the values S1-00 to S1-07 of the reference signals GIof the frequencies extracted in the signal extractor 90 and thecoefficients ref00 to ref07 are complex-multiplied for the frequencies,respectively. The coefficients ref00 to ref07 are conjugate complexnumbers of the known reference signal, and by performing the complexmultiplication, error vectors S2-00 to S2-07 with the periodicallyreceived reference signal are obtained. By previously adding a weightingcoefficient to the coefficients, the magnitudes of the error vectors canbe made the same.

FIG. 9 is a view showing the structure of the representative vectorcalculator 92. The representative vector calculator 92 has two complexadders 920-L and 920-H. In the representative vector calculator 92, theerror vectors S2-00 to S2-07 outputted from the error vector calculator91 are divided into two groups of low frequency and high frequency, andcomplex addition is performed for the groups by the complex adders 920-Land 920-H. Thereby, representative vectors S3L and S3H of the two groupsof low frequency and high frequency are obtained. The representativevectors which indicate the mean values of the groups may be divided bythe numbers of inputs of the complex adders. However, since what isimportant is the phases of the representative vectors, they are onlynecessarily positive real multiples of the mean value and it is notspecifically necessary to perform the division.

Now, using FIG. 10 to FIG. 13, the behaviors of the signal values whenthe SNR of the received signal is low will be described. FIG. 10 to FIG.13 show examples of the operations in the error vector calculator 91 andthe representative vector calculator 92.

FIG. 10 is a view showing the phase errors due to the residual carrierfrequency offset and the residual symbol synchronization shift in thefrequency domain in the case of a low SNR. In FIG. 10, the horizontalaxis shows the frequency, and the vertical axis shows the phase error inthe range of −π to π [rad]. Examples of the phase errors of the symbolsof the received signal calculated in the error vector calculator 91 areshown by the black circles.

In FIG. 10, in the phases (5), (7) and (8), phase discontinuity occursbecause the phase error due to the residual carrier frequency offset andthe residual symbol synchronization shift becomes large and the phasedisturbance due to the low SNR becomes large. When linear approximationis performed on these discontinuous phases by using the LSM, theobtained straight line is significantly different from the actual phaseerrors.

FIG. 11 is a view showing the error vectors of the phase errors in thecase of the low SNR shown in FIG. 10. FIG. 11 shows the error vectorsrepresented on the complex IQ plane. According to predeterminedfrequency ranges, the phases (1) to (4) are grouped into the lowfrequency vector and the phases (5) to (8), into the high frequencyvector.

FIG. 12 is a view showing the representative vectors of the errorvectors of the phase errors shown in FIG. 11. FIG. 12 shows therepresentative vectors of the error vectors represented on the complexIQ plane like FIG. 11. By the complex adder 920-L of the representativevector calculator 92, a low frequency representative vector S3L obtainedby the vector mean of the error vectors (1) to (4) is calculated.Moreover, by the complex adder 920-H, a high frequency representativevector S3H obtained by the vector mean of the error vectors (5) to (8)is calculated. To make the view more visible, the quotients when theoutputs of the complex adders 920-L and 920H are divided by the numbersof inputs of the complex adders are shown as the representative vectors.It is apparent that excellent mean values are obtained for the highfrequency representative vector S3H of the error vectors (5) to (8) byusing not phase calculation but vector mean calculation as describedabove.

FIG. 13 is a view showing the phase errors of the low frequencyrepresentative vector and the high frequency representative vector shownin FIG. 12. In FIG. 13, the horizontal axis shows the frequency, and thevertical axis shows the phase error in the range of −π to π [rad].

As shown in FIG. 13, compared with when phase discontinuity isdetermined based on the phase errors of the phases (1) to (8) of FIG.10, determination errors can be made few when the determination is madebased on two phases of the phase S3PL of the low frequencyrepresentative vector and the phase S3PH of the high frequencyrepresentative vector. This is because by obtaining the representativevector of each group by the mean, the number of times of determinationis reduced and the influence of the phase noise at a low SNR ismitigated by the mean calculation.

Next, the structure and operation of the correction value calculator 93will be described. FIG. 14 is a view showing the structure of thecorrection value calculator 93. The correction value calculator 93 hasvector-phase converter (vector to phase) 930-L and 930-H, a phaseinclination calculator 931, a phase offset calculator 932 and afrequency-by-frequency correction value calculator 933. The correctionvalue calculator 93 obtains the phase error correction value of eachfrequency from the representative vector obtained by the representativevector calculator 92.

In the correction value calculator 93, the low frequency representativevector S3L and the high frequency representative vector S3H areconverted into phases by the vector-phase converters 930-L and 930-H,respectively. The vector-phase conversion can be realized, for example,by an arc tan calculation or the CORDIC.

Then, by the phase inclination calculator 931, a phase error inclinationS4 a is obtained from the phase errors of the two representativevectors. The phase inclination calculator 931 has a wrapping subtractor9310 and a gain multiplicator 9311. In the wrapping subtractor 9310,wrapping subtraction is performed for the phases of the tworepresentative vectors, and the phase difference between therepresentative vectors is obtained in the range of −π to π [rad].

In the gain multiplicator 9311, multiplication by the gain which is thereciprocal of the quantity (not necessarily an integer) that is thefrequency difference between the representative vectors represented by afrequency number is performed. The gain which is the multiplier in thegain multiplicator 9311 is, for example, 1/32 which is the reciprocal ofthe difference between −16 which is the mean of frequency numbers −25,−22, −10 and −7 in the low frequency domain selected in FIGS. 7 and 16which is the mean of frequency numbers 8, 13, 19 and 24 in the highfrequency domain. In this case, the phase difference of the output ofthe wrapping subtractor 9310 is multiplied by the gain 1/32. By the gainmultiplication, the representative vector phase difference betweengroups is divided by the frequency difference, and the phase errorinclination S4 a is obtained.

Moreover, by the phase offset calculator 932, a phase error offset S4 bis obtained from the phase errors of the two representative vectors. Thephase offset calculator 932 has an adder 9320, a gain multiplier 9321, awrapping adder 9322, a selector 9323 and a determiner 9324.

In the adder 9320, the phases of the two representative vectors areadded together, and in the gain multiplicator 9321, the result of theaddition by the adder 9320 is multiplied by ½. The output of the gainmultiplicator 9321 is inputted to one input of the selector 9323 as itis, and a value which is the result of the wrapping addition of π by thewrapping adder 9322 is inputted to the other input of the selector 9323.In the determiner 9324, whether phase discontinuity is present or absentis determined. In the selector 9323, according to the result of thedetermination by the determiner 9324, one of the inputs is outputted asthe phase error offset S4 b. That is, when it is determined by thedeterminer 9324 that the phases are not discontinuous, the output of thegain multiplicator 9321 is outputted as it is from the selector 9323,and when it is determined by the determiner 9324 that the phases arediscontinuous, a value which is the output of the gain multiplicator9321 shifted by π is outputted from the selector 9323.

The phase discontinuity indicates that an excess phase of ±2π iscontained in any of the representative vectors. For this reason, withrespect to the result of addition of the phases of the representativevectors, the phase discontinuity can be corrected by shifting by π(shifting by −π is the same) after the multiplication by ½. Therefore,the phase error offset S4 b can be precisely calculated in the phaseoffset calculator 932.

Here, the wrapping subtraction by the wrapping subtractor 9310 and thewrapping addition by the wrapping adder 9322 will be described. Sincewrapping subtraction and wrapping addition are the same with respect tothe wrapping calculation although they are different in that one is sumand the other is difference, they will be collectively referred to aswrapping addition/subtraction in the following description.

FIG. 15 is a view showing the relationship between normaladdition/subtraction and wrapping addition/subtraction. The wrappingaddition/subtraction is to add an integral multiple of 2π when theaddition/subtraction value exceeds the range of ±π so that thecalculation result falls within the range of ±π. In the actualcalculation circuit, this can be easily realized, for example, byassociating the range of ±π with the range of a signed binaryinteger±((N−1)-th power of 2) and performing a calculation to extractthe lower N bits of the result of the addition/subtraction.

As a concrete example of the wrapping addition/subtraction, a case where±π is expressed by four bits will be shown. The MSB (Most SignificantBit) of the four bits represents whether the value is positive ornegative, and ±π is expressed by −8 to +7. FIG. 16 is a view showing thecorrespondence between the phases and the numerical values when thephases of ±π are assigned to four-bit values. As shown in FIG. 16, thephases of −π to π are assigned to integral values of −8 to +7 in π/8steps, and the integral values are expressed as binary values. Thebinary expressions of the integral values are 0→0000, +1→0001, +2→0010,+3→0011, +4→0100, +5→0101, +6→0110, +7→0111, −8→1000, −7→1001, −6→1010,−5→1011, −4→1100, −3→1101, −2→1110, and −1→1111.

When wrapping addition of −1 and −3 is performed, since −1 is 1111 and−3 is 1101 in binary expression and as shown by Expression [2] shownbelow, a carry occurs if normal binary addition is performed, the resultis 11100 which is five bits. In this example, the lower four bits areextracted from the five bits to obtain 1100, that is, −4, and −4 isobtained as the result of the wrapping addition of −1 and −3.

When wrapping addition of −6 and −3 is performed, although the result is−9 in the normal addition, since the range of ±π is exceeded, the resultof the wrapping addition is +7 from FIG. 15 and FIG. 16. In binaryexpression, −6 is 1010 and −3 is 1101 and as shown by Expression [3]shown below, the result is 10111 which is five bits when normal binaryaddition is performed. Here, the lower four bits are extracted to obtain0111, that is, +7 as the result of the wrapping addition of −6 and −3.By this calculation, even when the addition/subtraction result exceedsthe range of ±π, the wrapping addition/subtraction to obtain a resultfalling within the range of ±π by a shift by 2π can be realized.

$\begin{matrix}\left\lbrack {{Expression}\mspace{14mu} 1} \right\rbrack & \; \\\begin{matrix}{\mspace{31mu} {1\; 1\; 1\; 1}} & \Leftrightarrow & {- 1} \\\underset{\_}{{+ \; 1}\; 1\mspace{11mu} 0\; 1} & \Leftrightarrow & {- 3} \\{1{1\; 1\; 0\; 0}} & \Leftrightarrow & {- 4}\end{matrix} & \lbrack 2\rbrack \\\left\lbrack {{Expression}\mspace{14mu} 2} \right\rbrack & \; \\\begin{matrix}{\mspace{31mu} {10\; 10}} & \Leftrightarrow & {- 6} \\\underset{\_}{{+ \; 1}\; 1\mspace{11mu} 0\; 1} & \Leftrightarrow & {- 3} \\{1{0\; 1\; 1\; 1}} & \Leftrightarrow & {+ 7}\end{matrix} & \lbrack 3\rbrack\end{matrix}$

FIG. 17 is a view showing the structure of the determiner 9324 of thephase offset calculator 932 in the correction value calculator 93. Thedeterminer 9324 has a subtractor 93240, inequality sign determiners93241 a and 93241 b and an OR circuit 93242. In the subtractor 93240,the phases S3PL and S3PH of the two representative vectors aresubtracted. In the inequality sign determiner 93241 a, it is determinedwhether or not the result of the subtraction by the subtractor 93240 isnot less than π, that is, (S3PL−S3PH)≧π or not, and when the result ofthe determination is true, “1” is outputted. In the inequality signdeterminer 93241 b, whether or not the result of the subtraction by thesubtractor 93240 is less than −≧π, that is, (S3PL−S3PH)<−π or not, andwhen the result of the determination is true, “1” is outputted. In theOR circuit 93242, when any of the two outputs, that is, the outputs ofthe inequality sign determiners 93241 a and 93241 b is “1”, “1” isoutputted. The output of the OR circuit 93242 is a discontinuitydetermination value C1.

Thereby, when the absolute value of the calculated difference betweenthe phases of the two representative vectors is not less than π, thedeterminer 9324 determines that phase discontinuity is occurring, andoutputs “1” as the discontinuity determination value C1. Under acondition where symbol synchronization of the received signal isachieved, when no phase discontinuity is occurring, the phase error isnever π or larger. For this reason, whether phase discontinuity occursdue to the influence of the phase noise or not can be determined whetheror not the absolute value of the phase difference is not less than π.

FIG. 18 is a view showing the structure of the frequency-by-frequencycorrection value calculator 933 in the correction value calculator 93.In the frequency-by-frequency correction value calculator 933, from thephase error linearity shown in FIG. 3 and Expression [1], the phasecorrection values of the frequency numbers are obtained from the phaseerror inclination S4 a and the phase error offset S4 b.

The frequency-by-frequency correction value calculator 933 hasmultipliers 9330-00 to 9330-63 and adders 9331-00 to 9331-63. Here,since the frequency-by-frequency correction value is calculated for 64(64 symbols of the original GI) frequencies, 64 systems of circuits areprovided in parallel.

In the multipliers 9330-00 to 9330-63, in order to calculate the phaseerror correction value of each frequency number, the phase errorinclination S4 a obtained in the phase inclination calculator 931 ismultiplied by a coefficient corresponding to each frequency. Thecoefficients of the multiplication are frequency numbers −32 to +31. Inthe adders 9331-00 to 9331-63, the phase error offset S4 b obtained inthe phase offset calculator 932 is added to the results of themultiplication by the multipliers 9330-00 to 9330-63. By thesecalculations, frequency-by-frequency correction values S5-00 to S5-63are calculated.

FIG. 19 is a view showing the structure of the phase corrector 94. Thephase corrector 94 has phase-vector (phase to vector) convertors 940-00to 940-63, conjugate (conj) convertors 941-00 to 941-63 and complexmultipliers 942-00 to 942-63. Here, since the residual symbolsynchronization shift is corrected for 64 (64 symbols of the originalGI) frequencies, 64 systems of circuits are provided in parallel.

In the phase-vector converters 940-00 to 940-63, the phase errorcorrection values (frequency-by-frequency correction values) S5-00 toS5-63 of the frequencies obtained in the correction value calculator 93are converted into complex vectors. In the conjugate converters 941-00to 941-63, the complex vectors of the frequency-by-frequency correctionvalues are converted into conjugate complex numbers. In the complexmultipliers 942-00 to 942-63, received signals S0-00 to S0-63 in thefrequency domain having undergone transmission path error correction bythe transmission path corrector 8 are multiplied by the conjugatecomplex vectors of the frequency-by-frequency correction values.Thereby, the phases of the received signals are reversed, the phaseerrors are corrected, and signals S6-00 to S6-63 having undergone thecorrection are obtained. The phase-vector conversion can be realized by,for example, a tangent calculation or the CORDIC.

To the phase corrector 94, not only the structure where multiplicationby the conjugate complex vectors of the frequency-by-frequencycorrection values is performed but also various structures such as astructure where the phase is rotated by a gain multiplicator and aCORDIC section and a structure where phase correction is made by using acorrection table may be applied.

FIG. 20 is a view showing an example of the result of the phase errorestimation by a phase error corrector of the present embodiment. FIG. 20shows the result of a simulation of the symbol estimation error obtainedat the time of phase error correction when the symbol synchronizationshift is, for example, 0.2 symbols because of a shift of the samplingfrequency of the ADC, and is a view where the performances by the phaseerror estimation methods of the present embodiment and the conventionalexample are compared with each other.

In the graph of FIG. 20, the present embodiment is shown by O and theconventional example is shown by X; as the points representative of thevalues of the symbol estimation errors become lower in the figure, theerrors indicated thereby become smaller and highly precise correctioncan be made. In particular, it is apparent that the symbol estimationerror is extremely small compared with the conventional exampleparticularly when the CNR is as low as not more than 2 dB.

In the above-described embodiment, an example is described where inorder to obtain the low frequency representative vector and the highfrequency representative vector, four for each, a total of eightfrequencies are used. On the other hand, in the example of FIG. 20, theresult of a simulation is shown in a case where in order to use moreeffective measurement values, eight for each of the low frequency domainand the high frequency domain, a total of sixteen frequencies are used.

As described above, in the present embodiment, by using the periodicallyand repetitively transmitted specific reference signal (reference signalGI), by the phase error corrector 9, the specific reference signal(reference signal GI) is extracted from the received signal convertedinto the frequency domain and compared with the specific referencesignal to be transmitted, whereby the error vectors of the phase errorsare calculated in the frequency domain. A plurality of error vectors aredivided into not less than two error vector groups, the representativevalue of the error vectors is obtained for each group to calculate therepresentative vector, and the phase error inclination and offset arecalculated based on a plurality of representative vectors. Then, thephase error correction value of each frequency is calculated, and phaseerror correction for each frequency is performed. When therepresentative vector is obtained, the number of error vectors tocalculate the representative value in each group is necessarily at leasttwo, the number of groups into which the error vectors are divided isnecessarily at least two, and the phase error can be calculated based onat least four values.

For example, the phase error inclination is obtained from the differencebetween the phases of a plurality of representative vectors, and thephase error offset is obtained from the sum of the phases of therepresentative vectors. However, when the phase error offset isobtained, phase discontinuity is determined based on the differencebetween the phases of the representative vectors, and when the phasesare discontinuous, π is added to the sum of the phases of therepresentative vectors so that the result of the calculation fallswithin the range of ±π, and this is made the phase error offset.

Thereby, even when phase discontinuity occurs in the phase error, thephase discontinuity can be easily and precisely determined, and highlyprecise phase correction values can be obtained every time. Therefore,highly precise phase error correction can be realized by obtaininghighly precise phase correction values even in low SNR situations, and awireless communication system with high requirements for PER even at alow SNR and performing high-speed transmission such as the WiGig can behandled.

Application of the First Embodiment

FIG. 21 is a block diagram showing the structure where the phase errorcorrector of the present embodiment is applied to a receiver of awireless communication apparatus compliant with the OFDM. The wirelesscommunication apparatus shown in FIG. 21 has an RF processor 101, an ADCsection 102, a synchronization detector 104, a frequency corrector 105,an S/P converter 106, a DFT section 107, a transmission path corrector108, a phase error corrector 109 and a demodulator 113.

Compared with the structure of the wireless communication apparatuscompliant with the WiGig in FIG. 1, the wireless communication apparatuscompliant with the OFDM in FIG. 21 is different in that the IDFT sectionis absent and a signal in the frequency domain is demodulated and thatthe reference signal is not the GI assigned to a specific time but apilot carrier assigned to a specific frequency. The rest of thestructure and operation is similar to that of the wireless communicationapparatus in FIG. 1.

As described above, also in the wireless communication apparatuscompliant with the OFDM, by applying the phase error corrector of thepresent embodiment, even when the phase error or the noise level ishigh, a highly precise correction value can be obtained, so that highlyprecise phase error correction can be realized.

Second Embodiment

FIG. 22 is a view showing the spectrum of the reference signal GI in thefrequency domain and the frequency number specifying the low frequencydomain and the high frequency domain according to a second embodiment ofthe present disclosure.

The second embodiment is an example where the number of extracted phaseerrors used for error vector calculation is changed. In the signalextractor 90, all the signals in the frequency domain are extracted asshown in FIG. 22, the error vectors are calculated in the error vectorcalculator 91, and then, in the representative vector calculator 92, therepresentative vectors S3L and S3H are calculated for the high frequencydomain and the low frequency domain, respectively.

However, in the error vector calculator 91, to the magnitude of thecoefficient ref used for the calculation, a large weight is assignedwhen the spectrum of the GI is large, a small weight is assigned whenthe spectrum of the GI is small, and a weight is assigned for eachfrequency according to the magnitude of the absolute value of thespectrum. Thereby, noise is suppressed in the representative vectorobtained by the error vector synthesis in the representative vectorcalculator 92.

In this case, in the correction value calculator 93, also when thefrequency number corresponding to the representative vector iscalculated by the gain multiplicator 9311 of the phase inclinationcalculator 931, a weight is assigned to the frequency averaged in eachfrequency domain according to the magnitude of the spectrum of the GI.

By thus obtaining the phase error by extracting signals in amultiplicity of frequency domains, a more precise phase correction valuecan be obtained.

Third Embodiment

FIG. 23 is a block diagram showing the structure of a phase errorcorrector in a third embodiment of the present disclosure. The thirdembodiment is an example where the grouping of the frequency domains forwhich the representative vector is calculated is changed andrepresentative vectors of three groups of a low frequency domain, amedium frequency domain and a high frequency domain are obtained.

The phase error corrector 9 has a representative vector calculator 92Athat calculates three representative vectors corresponding to the threegroups of the low frequency domain, the medium frequency domain and thehigh frequency domain and a correction value calculator 93A thatcalculates the correction value from the three representative vectors.The rest of the structure and operation is similar to that of the phaseerror corrector of the first embodiment shown in FIG. 6.

Using FIG. 24 to FIG. 26, an example of the operation of therepresentative vector calculator 92A in the third embodiment when theSNR of the received signal is low will be described.

FIG. 24 is a view showing the phase errors due to the residual carrierfrequency offset and the residual symbol synchronization shift in thefrequency domain in the case of a low SNR. In FIG. 24, the horizontalaxis shows the frequency, and the vertical axis shows the phase error inthe range of −π to π [rad]. Examples of the phase errors of the symbolsof the received signal calculated in the error vector calculator 91 areshown by the black circles. In the present example, the low frequencydomain is phases (1) to (3), the medium frequency domain is phases (4)to (5), and the high frequency domain is phases (6) to (8). The valuesof the phase errors of the frequencies are similar to those of the firstembodiment shown in FIG. 10.

FIG. 25 is a view showing the error vectors of the phase errors in thecase of the low SNR shown in FIG. 24. FIG. 25 shows the error vectorsrepresented on the complex IQ plane. According to predeterminedfrequency ranges, the phases (1) to (3) are grouped into the lowfrequency vector, the phases (4) and (5), into the medium frequencyvector, and the phases (6) to (8), into the high frequency vector.

FIG. 26 is a view showing the representative vectors of the errorvectors of the phase errors shown in FIG. 25. By the representativevector calculator 92A, the low frequency representative vector S3L iscalculated by the vector mean of the error vectors (1) to (3), a mediumfrequency vector S3M is calculated by the vector mean of the errorvectors (4) and (5), and the high frequency representative vector S3H iscalculated by the vector mean of the error vectors (6) to (8).

FIG. 27 is a view showing the structure of the correction valuecalculator 93A in the third embodiment. The correction value calculator93A has vector-phase converters 930-L, 930-M and 930-H, phaseinclination calculators 931-LM and 931-MH, phase offset calculators932-LM and 932-MH, a phase inclination averaging section 934, a phaseoffset averaging section 935 and the frequency-by-frequency correctionvalue calculator 933.

In the correction value calculator 93A, the low frequency representativevector S3L is converted into a phase by the vector-phase converter930-L, the medium frequency representative vector S3M is converted intoa phase by the vector-phase converter 930-M, and the high frequencyrepresentative vector S3H is converted into a phase by the vector-phaseconverter 930-H.

Then, the inclination of the phase error between the low frequency andthe medium frequency is calculated from the phase errors of therepresentative vectors S3L and S3M by the phase inclination calculator931-LM, and the inclination of the phase error between the mediumfrequency and the high frequency is calculated from the phase errors ofthe representative vectors S3M and S3H by the phase inclinationcalculator 931-MH. The phase inclination calculators 931-LM and 931-MHare the same as the phase inclination calculator 931 of the firstembodiment.

Then, by the phase inclination averaging section 934, the mean of thetwo inclinations of the phase errors between the low frequency and themedium frequency and between the medium frequency and the high frequencyis calculated to obtain the phase error inclination S4 a. The phaseinclination averaging section 934 has an adder 9340 and a gainmultiplicator 9341. In the adder 9340, the inclinations of the phaseerrors between two different frequency domains are added together, andin the gain multiplicator 9341, the result of the addition by the adder9340 is multiplied by ½ to thereby perform mean calculation of the phaseerror inclinations. The output of the phase inclination averagingsection 934 is the phase error inclination S4 a obtained from the threerepresentative vectors of the low frequency domain, the medium frequencydomain and the high frequency domain.

Moreover, the phase error offset at the low frequency and the mediumfrequency is calculated from the phase errors of the representativevectors S3L and S3M by the phase offset calculator 932-LM, and the phaseerror offset at the low frequency and the medium frequency is calculatedfrom the phase errors of the representative vectors S3M and S3H by thephase offset calculator 932-MH. The phase offset calculators 932-LM and932-MH are the same as the phase offset calculator 932 of the firstembodiment.

Then, by the phase offset averaging section 935, the mean of the offsetsof the two phase errors between the low frequency and the mediumfrequency and between the medium frequency and the high frequency iscalculated to obtain the phase error offset S4 b. The phase offsetaveraging section 935 is the same as the phase offset calculator 932 ofthe first embodiment, and has an adder 9350, a gain multiplicator 9351,a wrapping adder 9352, a selector 9353 and a determiner 9354.

That is, in the phase offset averaging section 935, mean calculation ofthe phase error offset is performed by the adder 9350 and the gainmultiplicator 9351, and in the determiner 9354, whether phasediscontinuity is present or absent is determined. When it is determinedthat the phases are not discontinuous, the output of the gainmultiplicator 9351 is outputted as it is from the selector 9353, andwhen it is determined that the phases are discontinuous, a value whichis the output of the gain multiplicator 9351 shifted by it is outputtedfrom the selector 9353. The output of the phase offset averaging section935 is the phase error inclination offset S4 b obtained from the threerepresentative vectors of the low frequency domain, the medium frequencydomain and the high frequency domain.

The frequency-by-frequency correction value calculator 933 is similar tothat of the first embodiment, and from the linearity of the phase error,obtains the phase correction value of each frequency number from thephase error inclination S4 a and the phase error offset S4 b.

As described above, in the third embodiment, the error vectors aredivided into the three groups of the low frequency domain, the mediumfrequency domain and the high frequency domain, the representativevector of the error vectors is calculated for each group, and the phaseerror inclination and offset are calculated by the three representativevectors. Thereby, as in the first embodiment, a highly precise phasecorrection value can be obtained even in low SNR situations, so thathighly precise phase error correction can be realized.

The phase error correction value can also be calculated similarly to theabove when the error vectors are divided into not less than four groupsand the calculation of the representative vector for each group and thecalculation of the phase error inclination and offset are performed.

Fourth Embodiment

FIG. 28 is a block diagram showing the structure of a correction valuecalculator in a fourth embodiment of the present disclosure. The fourthembodiment is a modification of the correction value calculator 93A inthe above-described third embodiment. In the fourth embodiment, anexample is shown where the LSM is used for the calculation of the phaseerror inclination and offset.

The correction value calculator 93A of the fourth embodiment has thevector-phase converters 930-L, 930-M and 930-H, a phase unwrappingsection 938, an LSM approximator 939 and the frequency-by-frequencycorrection value calculator 933.

In the vector-phase converters 930-L, 930-M and 930-H, vector-phaseconversion of the low frequency representative vector S3L, the mediumfrequency representative vector S3M and the high frequencyrepresentative vector S3H is performed, respectively, and the phaseerrors of the representative vectors are obtained.

In the phase unwrapping section 938, for the phase error of eachrepresentative vector, phase unwrapping processing is performed, andwhen needed, an appropriate integral multiple of 2π is added to returnthe phase so that the absolute value of the difference between adjoiningphases does not exceed π. The phase unwrapping processing can berealized by using a generally known method by phase calculation.

In the LSM approximator 939, linear approximation by the LSM processingis performed, and from the phase errors of the representative vectorshaving continuity, the phase error inclination S4 a and the phase erroroffset S4 b are calculated. The LSM processing can be realized by usinga generally known method by approximation. In the frequency-by-frequencycorrection value calculator 933, from the linearity of the phase error,the phase correction value of each frequency number is obtained from thephase error inclination S4 a and the phase error offset S4 b.

As described above, a highly precise phase correction value can beobtained from the three representative vectors also by the method usingthe LSM.

Fifth Embodiment

FIG. 29 is a block diagram showing the structure of a receiver of awireless communication apparatus in a fifth embodiment of the presentdisclosure. The wireless communication apparatus of the fifth embodimentis a structure example where the structure of the wireless communicationapparatus compliant with the WiGig shown in FIG. 1 is partly changed andthe function of the phase corrector 94 of the phase error corrector 9 isunified into the transmission path corrector 8.

In the wireless communication apparatus of the fifth embodiment, thefunction of the phase error corrector 9 is divided into the phase errorestimator 95 and the phase corrector 94 and instead of the phase errorcorrector 9, the phase error estimator 95 and a transmission pathcorrector 8A having the function of the phase corrector 94 are provided.

FIG. 30 is a block diagram showing the structure of the phase errorcorrector in the fifth embodiment. The phase error estimator 95connected between the transmission path corrector 8A and the IDFTsection 10 has the signal extractor 90, the error vector calculator 91,the representative vector calculator 92 and the correction valuecalculator 93. That is, the phase error estimator 95 has a structurewhere the phase corrector 94 is removed from the phase error corrector 9shown in FIG. 6. Since the structures and operations of the elements aresimilar to those of the first embodiment, descriptions thereof areomitted. The phase correction value of each frequency calculated in thephase error estimator 95 is inputted to the transmission path corrector8A.

FIG. 31 is a view showing the structure of the transmission pathcorrector 8A in the fifth embodiment. In the transmission path corrector8A, the amplitude and the phase which are transmission characteristicspossessed by the transmission path between the transmitter and thereceiver are calculated, and the transmission path error is corrected.Since phase correction can be made on the phase component obtained bythe phase error estimator 95 together with the phases of thetransmission characteristics of the transmission path in thetransmission path corrector 8A, correction circuits can be reduced. Thepresent embodiment is a method where phase error correction by feedbackis performed unlike the phase error correction by feedforward shown inFIG. 6.

The transmission path corrector 8A has a transmission path estimator 80,multipliers 81-00 to 81-63, rotators 82-00 to 82-63, wrapping adders83-00 to 83-63, flip-flop circuits 84-00 to 84-63 with enable, andwrapping adders 85-00 to 85-63. Here, since amplitude and phasecorrection is performed for 64 (64 symbols of the original GI)frequencies, 64 systems of circuits are provided in parallel.

In the transmission path corrector 8A, by using the amplitude correctionvalue and the phase correction value obtained by the transmission pathestimator 80, the phase correction values of the frequencies calculatedby the phase error estimator 95 are synthesized, and amplitudecorrection and phase correction regarding the transmission pathcharacteristics are executed while phase error correction is performed.

In the transmission path estimator 80, transmission path estimation isperformed by using the output signal of the frequency corrector 5 or theP/S converter 11, and the amplitude correction value and the phasecorrection value according to the transmission characteristics possessedby the transmission path between the transmitter and the receiver arecalculated.

In the multipliers 81-00 to 81-63, the received signals S0-00 to S0-63in the frequency domain outputted from the DFT section 7 are multipliedby the amplitude correction value obtained in the transmission pathestimator 80 to perform amplitude correction. Moreover, in the rotators82-00 to 82-63, the phases of the outputs of the multipliers 81-00 to81-63 are rotated by the unified phase correction values outputted fromthe wrapping adders 85-00 to 85-63 to perform phase correction.

When this is done, in the wrapping adders 83-00 to 83-63, wrappingaddition of the correction values (phase error estimation outputs) S5-00to S5-63 of the phase errors of the frequencies outputted from the phaseerror estimator 95, to the outputs of the flip-flop circuits 84-00 to84-63 is performed. In the flip-flop circuits 84-00 to 84-63 withenable, the enable terminal holds the outputs of the wrapping adders83-00 to 83-63 at the time of high level. The flip-flop circuits 84-00to 84-63 are enabled at the time of the input of the received signal ofthe GI.

By these wrapping adders 83-00 to 83-63 and flip-flop circuits 84-00 to84-63, phases are accumulated every time the correction values S5-00 toS5-63 of the phase errors obtained from the phase error estimator 95 arechanged. The phase error change occurs when the received signal containsthe GI.

In the wrapping adders 85-00 to 85-63, wrapping addition of the outputsof the flip-flop circuits 84-00 to 84-63 and the phase correction valueobtained in the transmission path estimator 80 is performed. Thereby,the result of the phase error accumulation and the phase correctionvalue by the transmission path estimation are synthesized and inputtedto the rotators 82-00 to 82-63 to perform phase correction.

As described above, in the fifth embodiment, by unifying the phasecorrector for phase error correction and the phase corrector fortransmission path correction, correction circuits can be reduced, sothat the circuit scale can be reduced. Moreover, as in the firstembodiment, a highly precise phase correction value can be obtained evenin low SNR situations, so that highly precise phase error correction canbe realized.

Circumstances Leading up to the Contents of Another Embodiment of thePresent Disclosure

For example, the conventional examples of the phase error correctionmethods shown in the above-described Patent Documents 2 and 3 areexamples where in order to correct the residual carrier frequency offsetand the residual symbol synchronization shift, the residual carrierfrequency offset is obtained by a carrier frequency offset estimator andthen, the correction value is fed back to the phase error corrector toperform correction. Moreover, an example where the correction value isfed back to the frequency corrector to perform correction and an examplewhere the correction value is fed back to the RF processor to performcorrection are also present.

According to the method where the correction value is fed back toperform correction as in the above-described conventional example, sincethe correction of the residual carrier frequency offset is reflectedafter the feedback, when this is applied to a wireless communicationsystem that performs high-speed transmission, correction is sometimestoo late for the header of the received data. In particular, in the caseof the wireless communication standard WiGig (trademark; the sameapplies hereafter) (Wireless Gigabit) using a millimeter waveband, sincethe time of the preamble period is short, the correction of the residualcarrier frequency offset is not reflected from the header head of thereceived data, which causes a problem in that the header cannot becorrectly modulated. Therefore, a phase error correction method isdesired that enables correct demodulation of the received data byappropriately correcting the carrier frequency offset.

The present disclosure shows an example of a wireless communicationapparatus including DFT (Discrete Fourier Transformation) and IDFT(Inverse Discrete Fourier Transformation) in a receiver such as OFDM(Orthogonal Frequency Division Multiplexing) like a wireless LANstandard IEEE 802.11a,g,n or SC-FDE (Single Carrier Frequency DomainEqualizer) like the WiGig.

Correction of a phase error caused between a transmitter and a receiverdue to a residual carrier frequency offset and a residual symbolsynchronization shift in a wireless communication apparatus of this kindwill be described below.

FIG. 32 is a block diagram showing the structure of a receiver of awireless communication apparatus compliant with the WiGig. The wirelesscommunication apparatus shown in FIG. 32 has an RF (radio frequency)processor 1001, an ADC (Analog-Digital Converter) section 1002, an AGC(Auto Gain Controller) section 1003, a synchronization detector 1004, afrequency corrector 1005, an S/P (serial-parallel) converter 1006, a DFTsection 1007, a transmission path corrector 1008, a phase errorcorrector 1009, an IDFT section 1010, a P/S (parallel-serial) converter1011 and a demodulator 1013.

The RF processor 1001 converts a received signal of a radio frequencyreceived by the antenna into a baseband signal which is a complexsignal. The ADC section 1002 periodically samples the baseband signalwhich is the complex signal and converts it into a digital complexbaseband signal. The AGC section 1003 controls the gain of the signalamplification in the RF processor 1001 so that the output signal levelof the RF processor 1001 is maintained constant.

The synchronization detector 1004 detects the known preamble signal forsynchronization (STF described later) from the complex baseband signal.The frequency corrector 1005 calculates the error of the carrierfrequency by using the known preamble signal (STF described later), andperforms rough carrier frequency offset correction. The S/P converter1006 converts the complex baseband signal which is a serial signal intoa parallel signal. The DFT section 1007 converts the complex basebandsignal in the time domain having undergone the rough carrier frequencyoffset correction, into a complex signal in the frequency domain afterrough symbol synchronization according to the timing of the preamblesignal detected by the synchronization detector 1004.

The transmission path corrector 1008 corrects the transmission patherror between the transmitter and the receiver by using the knownpreamble signal (CEF described later). The phase error corrector 1009corrects, by using the known reference signal (GI described later), theresidual phase error caused by the residual carrier frequency offset andthe residual symbol synchronization shift.

The IDFT section 1010 converts the phase-error-corrected signal in thefrequency domain outputted from the phase error corrector 1009, into acomplex baseband signal in the time domain. The P/S converter 1011converts the parallel signal which is the output of the IDFT section1010, into a serial signal. The demodulator 1013 demodulates a digitallymodulated signal by using the complex baseband signal converted into thetime domain by the IDFT section 1010.

FIG. 33 is a view showing a signal format of the WiGig. The signaltransmitted in a WiGig wireless communication system has, from the head,an STF (Short Training Field), a CEF (Channel Estimation Field), a GI(Guard Interval), a header (Header), . . . , and data portions (Data1,Data2, . . . ). Here, an STF and a CEF are provided as preamble signals.

The STF is the repetition of 17 times of a known preamble signal Ga (128symbols) used in the AGC section 1003, the synchronization detector 1004and the frequency corrector 1005 of FIG. 32. During the AGC period fromthe head of the STF, an AGC operation by the AGC section 1003 isperformed, and during the remaining rough CFO period, the calculation ofthe rough carrier frequency offset by the frequency corrector 1005 isperformed. The last one symbol of the STF is a synchronization detectionperiod when rough symbol synchronization is performed by the detectionof the preamble signal by the synchronization detector 1004.

The CEF is the repetition of nine times of known preamble signals Ga, Gb(128 symbols), −Ga and −Gb different from the above-described STF usedin the transmission path corrector 1008 of FIG. 32. Here, Ga and Gb areprescribed code strings.

The header contains information representative of the attributes of thetransmission data such as the modulation method and the number oftransmitted symbols. The data portions contain the data to betransmitted itself. The GI is a known reference signal different fromthe above-mentioned STF and CEF and repetitively inserted at regulartime intervals in the header and the data portions. The GI is used asthe residual CFO calculation period when the calculation of the residualcarrier frequency offset by the phase error corrector 1009 is performed.

Next, the residual carrier frequency offset and the residual symbolsynchronization shift corrected in the phase error corrector 1009 willbe described. The carrier frequency offset is a phase error causedbecause the carrier frequency used when the complex baseband signal isorthogonally modulated in the RF processor of the transmitter (notshown) and the carrier frequency used for the orthogonal modulation inthe RF processor 1001 of the receiver are subtly different.

The frequency corrector 1005 estimates the error of the carrierfrequency (rough carrier frequency offset) to perform correction, andsince an error is caused in the estimation of the carrier frequencyoffset because of the influence of the signal noise and the phase noiseof the carrier, the phase error remains. This is the residual carrierfrequency offset.

The residual symbol synchronization shift is a phase error causedbecause the sampling frequency of the DAC (Digital Analog Converter)section generating the complex baseband signal in the transmitter (notshown) and the sampling frequency of the ADC section 1002 of thereceiver are subtly different. Because of the sampling frequency errorbetween the transmitter and the receiver, even if phase error correctionis performed in the earliest period, the symbol synchronization shiftremains and accumulates with the lapse of time, and the symbol timingerror increases. For this reason, it is necessary to correct theresidual symbol synchronization shift while continuously updating thephase error correction value.

A method of calculating the rough carrier frequency offset in thefrequency corrector 1005 will be described. FIG. 34 is a view showingthe Ga mutual correlation peak in the STF. FIG. 34 shows the mutualcorrelation peak of each of the N-th Ga (Ga(N)), the (N+1)-th Ga(Ga(N+1)) and (N+2)-th Ga (Ga(N+2)) represented on the complex IQ plane.

Between the mutual correlation peaks of the N-th Ga and the (N+1)-th Ga,a phase difference is present because of the carrier frequency offset,and similarly, between the correlation peaks of the (N+1)-th Ga and the(N+2)-th Ga, a phase difference is present. By averaging these two phasedifferences, the mean phase difference which is the noise component per128 symbols that is rounded is calculated, and the phase difference persymbol is obtained. The phase difference per symbol is the rough carrierfrequency offset. The larger the number of means when the phasedifference is calculated is, the higher the precision of the obtainedrough carrier frequency offset is.

Conventionally, a method is adopted where in order to correct theresidual phase error, the calculated residual carrier frequency offsetis fed back to the phase error corrector to perform correction as shownin the above-described Patent Documents 2 and 3. FIG. 35 is a blockdiagram showing the structure of a receiver of a wireless communicationapparatus using the conventional residual carrier frequency offsetcorrection method. In this conventional example, by a carrier frequencyoffset estimator 1514, the residual carrier frequency offset iscalculated from the output of an IDFT section 1510, and the correctionvalue is fed back to a phase error corrector 1509 to perform phase errorcorrection. A structure is also available where the correction value isfed back to a frequency corrector 1505 or an RF processor 1501 toperform correction.

FIG. 36 is a view showing the carrier frequency offset correction timingin the conventional example. In the STF, the estimation (calculation) ofthe rough carrier frequency offset is performed, and after thecalculation of the correction value, rough carrier frequency offsetcorrection is performed. After the rough carrier frequency offsetcorrection, the residual carrier frequency offset is accumulated.Thereafter, the estimation (calculation) of the residual carrierfrequency offset is performed in the GI, and after the calculation ofthe correction value, the residual carrier frequency offset correctionis performed.

In the conventional example, since the correction of the residualcarrier frequency offset obtained by the first GI is reflected after thefeedback, the correction processing is sometimes too late for theheader. In particular, when the residual carrier frequency offset islarge, since there is no timing of the residual carrier frequency offsetcorrection during the period of the CEF, a problem arises in that theheader cannot be correctly demodulated. Factors that increase theresidual carrier frequency offset include a low SNR (Signal-to-NoiseRatio), the phase noise of the carrier, and reduction in calculationprecision due to the number of means of Ga in the STF used for thecalculation of the rough carrier frequency offset being small.

In the wireless LAN standard IEEE 802.11a, the STF period is 8 μs,whereas in the WiGig, the STF period is as short as 1.236 μs. In thewireless communication apparatus compliant with the WiGig shown in FIG.32, for example, the RF processor 1001, and the ADC section 1002 and thesubsequent elements are formed of different circuit chips. For thisreason, the loop of the RF processor 1001, the ADC section 1002 and theAGC section 1003 performing the AGC operation takes time to transmit andreceive signals and uses much of the 1.236 μs of the STF period. Sincethe number of means of Ga in the STF used for the calculation of therough carrier frequency offset is small because of this, the precisionof the rough carrier frequency offset calculation is low and phase noiseof the carrier is caused, so that the residual carrier frequency offsetis large.

As described above, particularly in the wireless communication systemsuch as the WiGig performing high-speed transmission, if the residualcarrier frequency offset correction by the feedback as in theconventional example is performed, a period when correction is notperformed in a predetermined time occurs because of the time requiredfor the phase error calculation. For this reason, the residual carrierfrequency offset correction is too late for the header head of thereceived data, so that the header cannot be correctly demodulated.

In view of the above-mentioned problem, the present disclosure providesa reception apparatus and a phase error correction method and apparatuswhere the carrier frequency offset can be appropriately corrected evenat the header and the received data including the header can becorrectly demodulated.

Other Embodiments of the Present Disclosure

Hereinafter, embodiments according to the present disclosure will bedescribed in detail with reference to the drawings. The receptionapparatus, the phase error correction method and the phase errorcorrection apparatus according to the present disclosure are implementedin wireless communication apparatuses of the embodiments. In thedrawings used in the following description, the same elements aredenoted by the same reference numerals and signs and overlappingdescriptions are omitted.

Sixth Embodiment

FIG. 37 is a block diagram showing the structure of a receiver of awireless communication apparatus in a sixth embodiment of the presentdisclosure. The wireless communication apparatus of the sixth embodimenthas the RF processor 1001, the ADC section 1002, the AGC section 1003,the synchronization detector 1004, the frequency corrector 1005, the S/Pconverter 1006, the DFT section 1007, the transmission path corrector1008, the phase error corrector 1009, the IDFT section 1010, the P/Sconverter 1011, a residual carrier frequency offset corrector 1012 andthe demodulator 1013.

The RF processor 1001 amplifies a received signal of a radio frequencyreceived by the antenna, and performs orthogonal modulation thereon toconvert it into a baseband signal. The baseband signal having undergonethe orthogonal modulation is a complex signal.

The ADC section 1002 periodically samples the signal having undergonethe orthogonal modulation in the RF processor 1001, and converts it intoa digital complex baseband signal.

The AGC section 1003 calculates the amplitude of the digital complexbaseband signal, and controls the gain of the signal amplification inthe RF processor 1001 so that the output signal level of the RFprocessor 1001 is maintained constant. The AGC operation is performedduring the period of the known preamble signal (STF).

The synchronization detector 1004 detects the known preamble signal forsynchronization (STF) from the complex baseband signal, and outputs atiming signal for synchronization. The preamble signal is used forwindow synchronization, that is, rough symbol synchronization of the DFTsection 1007.

The frequency corrector 1005 calculates the rough carrier frequencyoffset as the carrier frequency error by using the known preamble signal(STF), and outputs a complex baseband signal obtained by correcting therough carrier frequency offset.

The S/P converter 1006 which is a buffer for operating the DFT section1007 converts the complex baseband signal which is a serial signal intoa parallel signal. The DFT section 1007 which corresponds to an exampleof a time-frequency converter performs time-frequency conversionaccording to the timing of the STF detected by the synchronizationdetector 1004 with respect to the complex baseband signal in the timedomain having undergone the rough carrier frequency offset correction,and outputs a complex signal in the frequency domain.

The transmission path corrector 1008 calculates the amplitude and thephase which are transmission characteristics possessed by thetransmission path between the transmitter and the receiver, by using theknown preamble signal (CEF), and corrects the transmission path error.

The phase error corrector 1009 calculates the residual carrier frequencyoffset and the residual symbol synchronization shift by using theperiodically inserted known reference signal (GI) as the specificreference signal, and corrects the phase error due to the residualsymbol synchronization shift in the frequency domain.

The IDFT section 1010 which corresponds to an example of thefrequency-time converter performs frequency-time conversion of theoutput signal of the phase error corrector 1009, and converts it into acomplex baseband signal in the time domain.

The P/S converter 1011 converts the parallel signal which is the outputof the IDFT section 1010, into a serial signal.

The residual carrier frequency offset corrector 1012 corrects theresidual carrier frequency offset in the time domain by using theresidual carrier frequency offset estimation value calculated by thephase error corrector 1009.

The demodulator 1013 demodulates a digitally modulated signal by usingthe complex baseband signal converted into the time domain and havingundergone the residual phase error correction, and obtains the receiveddata.

In the above-described structure, the synchronization detector 1004, thefrequency corrector 1005, the S/P converter 1006, the DFT section 1007,the transmission path corrector 1008, the phase error corrector 1009,the IDFT section 1010, the P/S converter 1011, the residual carrierfrequency offset corrector 1012 and the demodulator 1013 can beimplemented as an information processing circuit including a processorand a memory, and the functions can be realized by operating a softwareprogram in the processor to execute predetermined processing.

In the present embodiment, in the phase error corrector 1009, theresidual carrier frequency offset is calculated from the received signalconverted into the frequency domain by the DFT section 1007 andcorrected by the transmission path corrector 1008. Then, the calculatedresidual carrier frequency offset estimation value is supplied to theresidual carrier frequency offset corrector 1012 by feedforward, andresidual carrier frequency offset correction is performed on thereceived signal converted into the time domain by the IDFT section 1010.

FIG. 38 is a block diagram showing the structure of the phase errorcorrector 1009 in the sixth embodiment. The phase error corrector 1009has a signal extractor 1090, an error vector calculator 1091, a phaseerror calculator 1092, a residual phase error calculator 1093, aresidual symbol synchronization shift calculator 1094 and a residualsymbol synchronization shift corrector 1095.

In the signal extractor 1090, in the frequency domain, the periodicallyand repetitively received reference signal (GI) (corresponding to anexample of the received reference signal) is extracted from the receivedsignal. In the error vector calculator 1091, the reference signalextracted from the received signal and the known reference signal (GI)to be transmitted (corresponding to an example of the transmittedreference signal) are compared, and a plurality of error vectors due tothe difference therebetween are calculated. In the phase errorcalculator 1092, the error vectors obtained in the error vectorcalculator 1091 are converted into phases, and phase errors arecalculated.

In the residual phase error calculator 1093, phase error estimation bylinear approximation is performed from the phase errors obtained in thephase error calculator 1092, and the phase error offset and the phaseerror inclination are calculated. Here, the phase error inclination iscalculated as the residual symbol synchronization shift, and the phaseerror offset, as the residual carrier frequency offset. In the residualsymbol synchronization shift calculator 1094, the phase error estimationvalue at each frequency is calculated from the phase error inclinationobtained in the residual phase error calculator 1093.

In the residual symbol synchronization shift corrector 1095, theresidual symbol synchronization shift of the frequency is corrected byusing the phase error estimation value calculated in the residual phaseerror calculator 1093.

The phase error offset obtained in the residual phase error calculator1093 corresponds to the residual carrier frequency offset, and thisresidual carrier frequency offset estimation value is supplied to theresidual carrier frequency offset corrector 1012.

Next, the operation of the phase error corrector 1009 in the presentembodiment will be described in more detail.

In the signal extractor 1090, the reference signal GI is extracted fromthe received signal, and the spectrum shown in FIG. 7 where thereference signal GI of 64 symbols is Fourier-transformed is obtained.Here, the frequency number is the number representative of eachfrequency where 27.5 MHz which is the quotient when 1.76 GHz (−880 MHzto +880 MHz) which is the symbol rate of the WiGig standard is dividedby 64 symbols is one unit.

Of the spectrum in the frequency domain, particularly, one with a highabsolute value is high in noise tolerance, and is hardly affected byphase noise. Therefore, here, as an example, a spectrum of apredetermined number (eight symbols in the illustrated example) indecreasing order of the absolute value of the amplitude is used as therepresentative value, and the frequency numbers −25, −22, −10 and −7,and 8, 13, 19 and 24 shown by black circles in FIG. 7 are furtherextracted.

FIG. 39 is a view showing the structure of the error vector calculator1091. The error vector calculator 1091 has complex multipliers 1910-00to 1910-07. Here, since the error vectors of the reference signals GIare calculated for eight frequencies, eight systems of circuits areprovided in parallel. To calculate the error vectors, the referencesignal extracted from the received signal and the known reference signalto be transmitted are compared by using the complex multipliers 1910-00to 1910-07.

The complex multipliers 1910-00 to 1910-07 are supplied withcoefficients ref00 to ref07 of the known reference signal serving as thereference, respectively, and the values S11-00 to S11-07 of thereference signals GI of the frequencies extracted in the signalextractor 90 and the coefficients ref00 to ref07 are complex-multipliedfor the frequencies, respectively. The coefficients ref00 to ref07 areconjugate complex numbers of the known reference signal, and byperforming the complex multiplication, error vectors S12-00 to S12-07with the periodically received reference signal are obtained. Bypreviously adding a weighting coefficient to the coefficients, themagnitudes of the error vectors can be made the same.

FIG. 40 is a view showing the structure of the phase error calculator1092. The phase error calculator 1092 has vector-phase converters1920-00 to 1920-07 and unwrapping sections 1921-00 to 1921-07. Here,since phase errors are calculated for eight frequencies, eight systemsof circuits are provided in parallel.

In the vector-phase converters 1920-00 to 1920-07, the error vectorsS12-00 to S12-07 obtained in the error vector calculator 1091 areconverted into phases. The vector-phase conversion can be realized, forexample, by the arc tan calculation or the CORDIC.

In the unwrapping sections 1921-00 to 1921-07, phase unwrappingprocessing is performed to calculate phase errors S13-00 to S13-07.Here, when the phase is 2π+α, it is prevented from appearing as −2π+α,and the phase expression range is increased such as 2π+α by returningthe phase. The same applies when the phase is −2π−α. The phaseunwrapping processing can be implemented by using a generally knownmethod by phase calculation.

In the residual phase error calculator 1093 in FIG. 38, linearapproximation is performed based on the phase errors obtained in thephase error calculator 1092, and a phase error offset S14 b and a phaseerror inclination S14 a are calculated. For the linear approximation,for example, the LSM (Least Squares Method) is used. The LSM processingcan be realized by using a generally known method by approximation.

FIG. 41 is a view showing the structure of the residual symbolsynchronization shift calculator 1094. The residual symbolsynchronization shift calculator 1094 has multipliers 1940-00 to1940-63. Here, since the residual symbol synchronization shiftestimation value is calculated for 64 (64 symbols of the original GI)frequencies, 64 systems of circuits are provided in parallel. From thelinearity of the phase error shown in FIG. 3, the phase error (residualsymbol synchronization shift estimation value) of each frequency isobtained from the phase error inclination.

In the multipliers 1940-00 to 1940-63, in order to calculate theresidual symbol synchronization shift estimation value of each frequencynumber, the phase error inclination S14 a obtained in the residual phaseerror calculator 1093 is multiplied by a coefficient corresponding toeach frequency. The coefficients of the multiplication are frequencynumbers −32 to +31. By this coefficient multiplication, the phase errorsS15-00 to S15-063 of the frequencies are calculated.

FIG. 42 is a view showing the structure of the residual symbolsynchronization shift corrector 1095. The residual symbolsynchronization shift corrector 1095 has phase-vector (phase to vector)convertors 1950-00 to 1950-63, conjugate (conj) convertors 1951-00 to1951-63 and complex multipliers 1952-00 to 1952-63. Here, since theresidual symbol synchronization shift is corrected for 64 (64 symbols ofthe original GI) frequencies, 64 systems of circuits are provided inparallel.

In the phase-vector converters 1950-00 to 1950-63, the phase errors(residual symbol synchronization shift estimation values) S15-00 toS15-63 of the frequencies obtained in the residual symbolsynchronization shift calculator 1094 are converted into complexvectors. In the conjugate converters 1951-00 to 1951-63, the complexvectors of the residual symbol synchronization shift estimation valuesare converted into conjugate complex numbers. In the complex multipliers1952-00 to 1952-63, received signals S10-00 to S10-63 in the frequencydomain having undergone transmission path error correction by thetransmission path corrector 1008 are multiplied by the conjugate complexvectors of the residual symbol synchronization shift estimation values.Thereby, the phases of the received signals are reversed, the residualsymbol synchronization shifts are corrected, and signals S16-00 toS16-63 having undergone the correction are obtained. The phase-vectorconversion can be realized by, for example, the tangent calculation orthe CORDIC.

FIG. 43 is a view showing the structure of the residual carrierfrequency offset corrector 1012. The residual carrier frequency offsetcorrector 1012 has a phase-vector (phase to vector) convertor 1120, aconjugate (conj) convertor 1121 and a complex multiplier 1122. Theresidual carrier frequency offset corrector 1012 performs phase errorcorrection in the time domain.

In the phase-vector converter 1120, the phase error offset S14 b, thatis, the residual carrier frequency offset estimation value obtained inthe residual phase error calculator 1093 is converted into a complexvector. In the conjugate converter 1121, the complex vector of theresidual carrier frequency offset estimation value is converted into aconjugate complex number. In the complex multiplier 1122, a receivedsignal S17 in the time domain is multiplied by the conjugate complexvector of the residual carrier frequency offset estimation value.Thereby, the phase of the received signal S17 in the time domain isrotated, the residual carrier frequency offset is corrected, and asignal S18 having undergone the correction is obtained. The phase-vectorconversion can be realized by, for example, the tangent calculation orthe CORDIC.

FIG. 44 is a view showing the carrier frequency offset correction timingin the present embodiment. In the STF, the estimation (calculation) ofthe rough carrier frequency offset is performed by the frequencycorrector 1005, and after the calculation of the correction value, roughcarrier frequency offset correction is performed. After the roughcarrier frequency offset correction, the residual carrier frequencyoffset is accumulated. In the CEF, transmission path error correction isperformed by the transmission path corrector 1008.

Thereafter, the estimation (calculation) of the residual carrierfrequency offset is performed by the phase error corrector 1009 in theGI in front of the header, and the residual carrier frequency offsetcorrection is performed by the residual carrier frequency offsetcorrector 1012.

In the present embodiment, by using the periodically and repetitivelytransmitted specific reference signal (reference signal GI), by thephase error corrector 1009, the specific reference signal (referencesignal GI) is extracted from the received signal converted into thefrequency domain and compared with the specific reference signal to betransmitted, whereby the phase error offset and the phase errorinclination are calculated in the frequency domain. Based on thecalculated phase error inclination, residual symbol synchronizationshift correction is performed in the frequency domain by the phase errorcorrector 9.

Moreover, the calculated phase error offset (residual carrier frequencyoffset estimation value) is supplied to the residual carrier frequencyoffset corrector 1012 as a phase rotation angle in the time domain, andthe phase of the received signal converted to the time domain isrotated. That is, residual carrier frequency offset correction isperformed in the time domain by the residual carrier frequency offsetcorrector 1012. Since the frequency-time conversion processing in theIDFT section 1010 takes time, the processing delay of the phase errorcorrection can be eliminated by feed-forwarding the residual carrierfrequency offset estimation value to the residual carrier frequencyoffset corrector 1012 and executing the residual carrier frequencyoffset correction in the time domain.

According to the present embodiment, even in a wireless communicationsystem performing high-speed transmission, residual carrier frequencyoffset can be executed from the head of the header immediatelysucceeding the GI, so that the header can be correctly demodulated inthe demodulator 1013. Consequently, even when the residual carrierfrequency offset is large, the received data including the header can becorrectly demodulated without the use of a signal delay buffer.

Seventh Embodiment

FIG. 45 is a block diagram showing the structure of a residual carrierfrequency offset corrector in a seventh embodiment of the presentdisclosure. The seventh embodiment is an example where the structure ofthe residual carrier frequency offset corrector 1012 is changed. Therest of the structure is similar to that of the sixth embodimentdescribed above.

The residual carrier frequency offset corrector 1012 of the seventhembodiment has a gain multiplicator 1123 and a CORDIC section 1124. Inthe gain multiplicator 1123, the phase error offset S14 b, that is, theresidual carrier frequency offset estimation value obtained in theresidual phase error calculator 1093 is multiplied by −1. In the CORDICsection 1124, the phase of the received signal S17 in the time domain isrotated by the residual carrier frequency offset estimation valuemultiplied by −1, and the residual carrier frequency offset iscorrected.

In the seventh embodiment, functions similar to those of the sixthembodiment can be realized, and the header can be correctly demodulatedalso in a wireless communication system performing high-speedtransmission.

Eighth Embodiment

FIG. 46 is a block diagram showing the structure of a residual symbolsynchronization shift corrector in an eighth embodiment of the presentdisclosure. The eighth embodiment is an example where the structure ofthe residual symbol synchronization shift corrector 1095 is changed. Therest of the structure is similar to that of the sixth embodimentdescribed above.

The residual symbol synchronization shift corrector 1095 of the eighthembodiment has gain multiplicators 1953-00 to 1953-63 and CORDICsections 1954-00 to 1954-63. In the gain multiplicators 1953-00 to1953-63, the phase errors (residual symbol synchronization shiftestimation values) S15-00 to S15-63 of the frequencies obtained in theresidual symbol synchronization shift calculator 1094 are multiplied by−1. In the CORDIC sections 1954-00 to 1954-63, the phases of thereceived signals S10-00 to S10-63 in the frequency domain are rotated bythe residual symbol synchronization shift estimation values of thefrequencies multiplied by −1, and the residual symbol synchronizationshifts are corrected.

In the eighth embodiment, functions similar to those of the sixthembodiment can be realized, and the header can be correctly demodulatedalso in a wireless communication system performing high-speedtransmission.

Ninth Embodiment

FIG. 47 is a block diagram showing the structure of a receiver of awireless communication apparatus in a ninth embodiment of the presentdisclosure. The wireless communication apparatus of the ninth embodimentis a structure example where a residual carrier frequency offsetcorrector 1012A is provided between the IDFT section 1010 and the P/Sconverter 1011. The rest of the structure is similar to that of thesixth embodiment described above.

In this case, the residual carrier frequency offset corrector 1012Acorrects the residual carrier frequency offsets of the received signalsin the time domain by 64 parallel circuits. The structure of theresidual carrier frequency offset corrector 1012A is such that 64residual carrier frequency offset correctors 1012 of the structure ofFIG. 43 or the structure of FIG. 45 are arranged in parallel.

In the ninth embodiment, functions similar to those of the sixthembodiment can be realized, and the header can be correctly demodulatedalso in a wireless communication system performing high-speedtransmission.

Tenth Embodiment

FIG. 48 is a block diagram showing the structure of a receiver of awireless communication apparatus in a tenth embodiment of the presentdisclosure. The wireless communication apparatus of the tenth embodimenthas a phase error estimator 1015 and a time domain residual symbolsynchronization shift corrector 1016 instead of the phase errorcorrector 1009 in the sixth embodiment shown in FIG. 37.

The phase error estimator 1015 is connected between the transmissionpath corrector 1008 and the IDFT section 1010, and estimates(calculates) the residual symbol synchronization shift estimation valueand the residual carrier frequency offset estimation value. The timedomain residual symbol synchronization shift corrector 1016 is providedbetween the P/S converter 1011 and the residual carrier frequency offsetcorrector 1012, and performs residual symbol synchronization shiftcorrection of the received signal in the time domain. The rest of thestructure is similar to that of the sixth embodiment described above.

In the tenth embodiment, from the received signal converted to thefrequency domain by the DFT section 1007, the residual symbolsynchronization shift estimation value and the residual carrierfrequency offset estimation value are calculated by the phase errorestimator 1015. The residual symbol synchronization shift estimationvalue is supplied to the time domain residual symbol synchronizationshift corrector 1016, the residual carrier frequency offset estimationvalue is supplied to the residual carrier frequency offset corrector1012, and on the received signal converted to the time domain by theIDFT section 1010, residual symbol synchronization shift correction andresidual carrier frequency offset correction are performed each in thetime domain.

FIG. 49 is a view showing the structure of the phase error estimator1015. The phase error estimator 1015 has the signal extractor 1090, theerror vector calculator 1091, the phase error calculator 1092 and theresidual phase error calculator 1093. That is, the phase error estimator1015 has a structure where the residual symbol synchronization shiftcalculator 1094 and the residual symbol synchronization shift corrector1095 are removed from the phase error corrector 1009 of the sixthembodiment. Therefore, detailed descriptions of the operations of theelements are omitted here.

In the phase error estimator 1015, for the reference signal GI extractedin the signal extractor 1090, the error vector is calculated in theerror vector calculator 1091 and converted into a phase error in thephase error calculator 1092, and the phase error offset S14 b and thephase error inclination S14 a are calculated in the residual phase errorcalculator 1093. The phase error inclination S14 a is outputted as theresidual symbol synchronization shift estimation value, and the phaseerror offset S14 b, as the residual carrier frequency offset estimationvalue.

FIG. 50 is a view showing the structure of the time domain residualsymbol synchronization shift corrector 1016. The time domain residualsymbol synchronization shift corrector 1016 performs, in order tocorrect the residual symbol synchronization shift in the time domain,not phase rotation but synchronization correction by using a filter. Thetime domain residual symbol synchronization shift corrector 1016 has anIQ separator 1160, a correction coefficient selector 1161, flip-flop(FF) sections 1162-I-00 to 1162-I-09 and 1162-Q-00 to 1162-Q-09,multipliers 1163-I-00 to 1163-I-10 and 1163-Q-00 to 1163-Q-10, adders1164-I and 1164-Q and an IQ unification section 1165. Here, an examplewhere the number of filter taps is 11 is shown as an example.

In the time domain residual symbol synchronization shift corrector 1016,in the IQ separator 1160, the complex signal of the received signal S10in the time domain is separated into quadrature components of I and Q.In the FF sections 1162-I-00 to 1162-I-09 and 1162-Q-00 to 1162-Q-09,the complex signal separated into I and Q is held. In the correctioncoefficient selector 1161, the correction coefficient of each tap of thefilter is selected from the phase error inclination S14 a. In themultipliers 1163-I-00 to 1163-I-10 and 1163-Q-00 to 1163-Q-10, theselected correction coefficients and the signals held in the FF sectionsare multiplied. In the adders 1164-I and 1164-Q, the multiplicationresults of the multipliers are added, and the sum of each of I and Q iscalculated. In the IQ unification section 1165, the addition results ofI and Q are unified and converted into a complex signal to obtain thesignal S17 having undergone the time domain symbol synchronization shiftcorrection.

FIG. 51 and FIG. 52 are views explaining the correction coefficientselection processing in the correction coefficient selector 1161. FIG.51 is a view showing the relationship between the phase errorinclination and the residual symbol synchronization shift correctionvalue, and FIG. 52 is a view showing an example of the correctioncoefficient selection. In FIG. 51, the horizontal axis shows the phaseerror inclination, and the vertical axis shows the residual symbolsynchronization shift correction value. The linear function 1166 shownin FIG. 51 represents residual symbol synchronization shift correctionvalues −1 to +1 corresponding one to one to phase error inclinations+π/64 to −π/64. For example, when the phase error inclination S14 a is avalue al shown at the black circle in the figure, the residual symbolsynchronization shift correction value corresponding to the linearfunction 1166 is a value cal shown at the white circle in the figure.The correction coefficient selector 1161 obtains the correspondingresidual symbol synchronization shift correction value from the phaseerror inclination.

Then, the correction coefficient selector 1161 selects the correctioncoefficient of each tap from the residual symbol synchronization shiftcorrection values corresponding to the phase error inclinations, forexample, by using the coefficient of the sinc function shown in FIG. 52as the tap coefficient. In FIG. 52, the horizontal axis shows the tapnumber corresponding to each tap, and the vertical axis shows the value(correction coefficient) of the sinc function.

In the example of FIG. 52, when the residual symbol synchronizationshift correction value is 0, since the symbol synchronization shift is0, a value ha0 at the black circle in the figure, that is, the valuewhere the center is “1” and the rest is “0” is selected as thecorrection coefficient. When the residual symbol synchronization shiftcorrection value is 0.2 (the symbol synchronization shift is −0.2), avalue (the value at the black square in the figure) ha2 shifted by +0.2(corresponding to two marks on the right in the figure) from thecorrection value 0 is selected as the correction coefficient.

By performing synchronization correction by using the above-describedfilter and passing the received signal in the time domain therethrough,the residual symbol synchronization shift in the time domain can becorrected. The filter coefficient may be adjusted according to theresidual symbol synchronization shift value or the like.

In the tenth embodiment, functions similar to those of the sixthembodiment can be realized, and the header can be correctly demodulatedalso in a wireless communication system performing high-speedtransmission.

In the tenth embodiment, by performing residual symbol synchronizationshift correction and residual carrier frequency offset correction in thetime domain, the residual symbol synchronization shift can also becorrected from the header head together with the residual carrierfrequency offset. Consequently, the correction of the phase error due tothe residual symbol synchronization shift can be performed earlier.

Eleventh Embodiment

FIG. 53 is a block diagram showing the structure of a receiver of awireless communication apparatus in an eleventh embodiment of thepresent disclosure. The wireless communication apparatus of the eleventhembodiment is an example where in the structure of the tenth embodiment,the order of the time domain residual symbol synchronization shiftcorrector 1016 and the residual carrier frequency offset corrector 1012is reversed and the residual carrier frequency offset corrector 1012 isdisposed in the preceding stage. The rest of the structure is similar tothat of the above-described tenth embodiment.

In the eleventh embodiment, functions similar to those of the tenthembodiment can be realized, and the header can be correctly demodulatedalso in a wireless communication system performing high-speedtransmission.

In the eleventh embodiment, by performing residual carrier frequencyoffset correction in advance in consideration of the calculation errorof the CORDIC section or the like in the residual carrier frequencyoffset corrector 1012, the precision of the residual symbolsynchronization shift correction can be improved.

Circumstances Leading up to the Contents of Still Another Embodiment ofthe Present Disclosure

One modulation method in the wireless communication is a modulationmethod where the phase is rotated (for example, π/2 shift BPSK [BinaryPhase Shift Keying] modulation, π/4 shift BPSK modulation).

As a method where the correlation value is obtained for a signal streammodulated by the modulation method where the phase is rotated, a methodis known where the phase of the received signal is reversely rotatedaccording to the rotation amount compliant with the modulation methodand the correlation value is obtained from the reversely rotatedreceived signal (for example, see Reference Patent Document 1).

(Reference Patent Document 1) Japanese Patent No. 3811002

In the reception apparatus of Reference Patent Document 1, when thecarrier frequency error between the transmission and receptionapparatuses is estimated by using the obtained correlation value, theprecision of the carrier frequency error estimation is insufficient if asampling frequency error is present between the transmission andreception apparatuses. Accordingly, a reception apparatus and areception method capable of improving the precision of the carrierfrequency error estimation is desired.

In the wireless communication, for example, the carrier frequency errorbetween the transmission and reception apparatuses is one factor thatdecreases the communication quality. The carrier frequency error is thedifference in frequency between the carrier wave used by thetransmission apparatus and the carrier wave used by the receiver.

In the wireless communication, there are cases where the carrierfrequency error between the transmission and reception apparatuses isestimated, for example, by using the maximum value of the correlationvalues of the repetition section (for example, the preamble in acommunication signal) of a known signal.

In the above-described reception apparatus of Reference Patent Document1, for example, a signal rotated by π/2 radians is received, thecorrelation values are calculated by reversely rotating the phase of thereceived signal by π/2 radians, and the maximum value of the correlationvalues is detected.

When the above-described reception apparatus of Reference PatentDocument 1 is applied to the estimation of the carrier frequency error,a correlation value obtained from a complex number is used. When the sumof the phase rotation amount by the π/2 shift of the transmissionapparatus and the phase reverse rotation amount by the −π/2 shift of thephase reverse rotator of the reception apparatus is fixed, the phase ofthe maximum value of the correlation values obtained in the repetitionsection is rotated according to the carrier frequency error between thetransmission and reception apparatuses.

FIG. 54 is a schematic view showing the manner of phase rotation of themaximum correlation value in the reception processing by the receptionapparatus of Reference Patent Document 1. In the reception apparatus ofReference Patent Document 1, when the phase of the maximum correlationvalue is rotated, the carrier frequency error can be estimated from themean phase rotation amount (the inclination of the straight lineconnecting the points in FIG. 55) of the maximum correlation value.

However, when a sampling frequency error is present between thetransmission and reception apparatuses, the timing of the −π/2 shiftwith respect to the π/2-shifted received signal is gradually shifted.

FIG. 55 is a schematic view showing a phase change of the maximumcorrelation value when a carrier frequency error and a samplingfrequency error are present between the transmission and receptionapparatuses. FIG. 55 shows a case where the transmission samplingfrequency is lower than the reception sampling frequency.

In FIG. 55, in parts 2190 surrounded by dotted lines, the phase of themaximum correlation value is rotated by approximately −π/2 radians. Thatis, discontinuous points are preset in the phase change of the maximumcorrelation value. Consequently, the mean phase rotation amount of themaximum correlation value is discontinuous, so that the precision of thecarrier frequency error estimation is deteriorated.

Hereinafter, a reception apparatus and a reception method capable ofimproving the precision of the carrier frequency error estimation willbe described. Here, some structure examples where the above-describedrough carrier frequency offset correction is performed are shown.

Yet Other Embodiments of the Present Disclosure Twelfth Embodiment

FIG. 56 is a block diagram showing a structure example of a receptionapparatus 2000 in a twelfth embodiment. The reception apparatus 2000 isprovided with a sampling section 2101, a phase reverse rotator 2103, acorrelation value calculator 2105, a phase rotator 2107, a maximum valuedetector 2109, a carrier frequency error estimator 2111 and a frequencycorrector 2113. FIG. 56 includes a part associated with the correctionof the carrier frequency error.

The reception apparatus 2000 receives a signal modulated by apredetermined modulation method from a transmission apparatus (notshown). The predetermined modulation method includes, for example, themodulation method where the phase is rotated. Moreover, the receptionapparatus 2000 successively receives received signals, and theprocessings in the succeeding blocks are successively performed.

The sampling section 2101 samples a received signal 2100 by apredetermined sampling frequency Fs_r, and outputs a received sample2102. Here, when the symbol rate of the received signal 2100 is Rsym,Fs_r≅Rsym as an example. That is, the sampling section 21011×-oversamples the received signal 2100. Here, the sampling frequency ofthe transmission apparatus is Fs_t, and the transmission apparatus alsoperforms 1×-oversampling. That is, Fs_t=Rsym.

The phase reverse rotator 2103 reversely rotates the phase of thereceived sample 2102 according to the rotation amount compliant with themodulation method, and outputs a received sample 2104 the phase of whichhas been rotated reversely. As the modulation method, for example, theπ/2 shift BPSK is used. When the modulation method is the π/2 shiftBPSK, the phase reverse rotator 2103 reversely rotates the phase of thereceived sample 2102 by −π/2 radians (shifts by −π/2) as shown byExpression [4]:

[Expression 3]

Rx′(n)=Rx(n)×exp(−1i×π/2×(n−1)), n=1,2, . . .   [4]

Here, Rx is the received sample 2102, and Rx′ is the −π/2-shiftedreceived sample 2104.

The correlation value calculator 2105 calculates a correlation value2106 of a predetermined known signal stream (for example, the signalstream used for the preamble) and a predetermined signal stream (forexample, the signal stream of the preamble) in the received sample 2104,and outputs the correlation value 2106.

For example, in a wireless LAN standard IEEE 802.11ad using a millimeterwave in the 60 GHz band, it is prescribed that π/2-shift-BPSK (BinaryPhase Shift Keying)-modulated Golay series is repetitively transmittedin the preamble section. The reception apparatus 2000 receives, forexample, a signal used for millimeter waves.

The phase rotator 2107 rotates the phase of the correlation value 2106successively derived from the rotation amount compliant with themodulation method, and outputs a phase-rotated correlation value 2108.For example, when the modulation method is the π/2 shift BPSK, the phaserotator 2107 rotates the phase of the correlation value 2106 by π/2radians (shifts by π/2) as shown by Expression [5].

[Expression 4]

C′(n)=C(n)×exp(1i×π/2×(n−1)), n=1,2, . . .   [5]

Here, C is the correlation value 2106 and C′ is the π/2-shiftedcorrelation value 2108.

FIG. 57 to FIG. 59 are schematic views showing structure examples of thephase rotator 2107. Since the correlation value 2106 and the correlationvalue 2108 are expressed by complex numbers, in FIG. 57 to FIG. 59, thecorrelation value 2106 and the correlation value 2108 are shown in astate of being divided into I components and Q components.

FIG. 57 is a schematic view showing a structure example in a case wherethe phase rotator 2107 includes complex multipliers. A cosine wavegenerator 2107_1 and a sinusoidal wave generator 2107_2 generate asignal that repeats a change every sample. This signal is expressed, forexample, by (cos, sin)=(1, 0), (0, 1), (−1, 0), (0, −1). In FIG. 57, thecosine wave generator 2107_1 is represented as “cos”, and the sinusoidalwave generator 2107_2 is represented by “sin”.

Moreover, multipliers 2107_3 a, 2107_3 b, 2107_3 c and 2107_3 d multiplya correlation value 2106 _(—) i and a correlation value 2106 _(—) q. Anadder 2107_5 a and an adder 2107_5 b add the multiplication result, andoutput a correlation value 2108 _(—) i and a correlation value 2108 _(—)q. Thereby, the phase of the correlation value 2106 is rotated by π/2radians.

FIG. 58 is a schematic view showing a structure example in a case wherethe phase rotator 2107 includes a counter and a selector and performsI/Q change and sign inversion. A counter 2107_7 counts every sample, forexample, counts as 0, 1, 2, 3, 0, . . .

On the signal lines connected to the selector numbers “1” and “3” of aselector 2107_9, I/Q change is performed on the correlation value 2106_(—) i and the correlation value 2106 _(—) q. Moreover, on the I signalline connected to the selector number “1”, the I signal line and the Qsignal line connected to the selector number “2” and the I signal lineconnected to the selector number “3”, sign inversion is performed by asign inverter.

The selector 2107_9 selects a signal according to the count value, forexample, selects 0, 1, 2, 3, 0, . . . , and outputs the correlationvalue 2108 _(—) i and the correlation value 2108 _(—) q. Thereby, thephase of the correlation value 2106 is rotated by π/2 radians.

In the structure of FIG. 58, since neither a multiplier nor an adder isused, the circuit size can be reduced compared with the structure ofFIG. 57, so that power consumption can be reduced.

FIG. 59 is a schematic view showing a structure example in a case wherethe phase rotator 2107 uses the CORDIC (Coordinate Rotational DigitalComputer) algorithm.

A CORDIC calculation circuit 2107_11 inputs a predetermined phase asphase information 2107_10. The predetermined phase is, for example, 0radians, π/2 radians, π radians, 3π/2 radians, 0 radians, . . .

The CORDIC calculation circuit 2107_11 generates the correlation value2108 _(—) i and the correlation value 2108 _(—) q from the predeterminedphase, the correlation value 2106 _(—) i and the correlation value 2106_(—) q and outputs them. Thereby, the phase of the correlation value2106 is rotated by π/2 radians.

In the structure of FIG. 59, since no multiplier is used, the circuitsize can be reduced compared with the structure of FIG. 57, so thatpower consumption can be reduced.

When the length of the known signal stream is L, the maximum valuedetector 2109 detects the correlation value (maximum correlation value)where the power in the L sample period is maximum, from among thesuccessively derived correlation values 2108, and outputs the maximumcorrelation value 2110. For example, in FIG. 60, when L=32, a pluralityof correlation values 2108 are sectioned at every 32 sample periods. Themaximum value detector 2109 detects the maximum correlation value 2110from among the correlation values in the sectioned periods. The L sampleperiod is an example of periods divided according to the length of theknown signal stream.

Now, using FIG. 61(A) to 61(C), FIG. 62(A) to FIG. 62(C) and FIG. 63(A)to FIG. 63(C), change of the characteristics of the maximum correlationvalue considering the carrier frequency error and the sampling frequencyerror will be described. In FIG. 61(A) to 61(C), FIG. 62(A) to FIG.62(C) and FIG. 63(A) to FIG. 63(C), it is assumed that the phase rotator2107 is absent.

FIG. 61(A) is a schematic view showing an example of the time change ofthe index of the maximum correction value in a case where Fs_t=Fs_r.FIG. 61(B) is a schematic view showing an example of the time change ofthe amplitude of the maximum correlation value in the case whereFs_t=Fs_r. FIG. 61(C) is a schematic view showing an example of the timechange of the phase of the maximum correlation value in the case whereFs_t=Fs_r.

The index of the maximum correlation value is a value representative ofthe position where the maximum value appears in the range sectioned bythe L sample period. For example, when L=32 and the maximum correlationvalue appears at the 26th one of the 32 samples, the index=26.

In FIG. 61(A) and FIG. 61(B), in the case where Fs_t=Fs_r, the index andamplitude of the maximum correlation value do not change even thoughtime elapses. On the other hand, in FIG. 61(C), the phase of the maximumcorrelation value is continuously rotated as time elapses because of theinfluence of the carrier frequency error. That is, the phase is rotatedwith a substantially fixed rotation amount.

FIG. 62(A) is a schematic view showing an example of the time change ofthe index of the maximum correction value in the conventional case whereFs_t<Fs_r. FIG. 62(B) is a schematic view showing an example of the timechange of the amplitude of the maximum correction value in the casewhere Fs_t<Fs_r. FIG. 62(C) is a schematic view showing an example ofthe time change of the phase of the maximum correction value in the casewhere Fs_t<Fs_r.

In the case where Fs_t<Fs_r, since the sample timing of the receivedsignal by the reception apparatus gradually advances with respect to thesample timing of the transmission apparatus, the index increases as timeelapses. That is, the position of appearance of the maximum correlationvalue is shifted to a sampling position in the rear.

Moreover, in a case where Fs_r<Rsym, if the sample timing is shifted,the symbol synchronization timing is shifted. In this case, theamplitude of the maximum correlation value becomes maximum at the pointof time when the symbol synchronization timings coincide in thetransmission apparatus and the reception apparatus.

On the other hand, the amplitude of the maximum correlation valuebecomes minimum at the point of time when the symbol synchronizationtiming is shifted by ½ symbol. Referring to FIG. 62(A) and FIG. 62(B),it can be understood that the index is changed when the amplitude of themaximum correlation value is minimum, that is, when the symbolsynchronization timing is shifted by ½ symbol.

The phase of the maximum correlation value is sometimes rotated by −π/2radians as in FIG. 55, and is discontinuously changed. When the index ischanged, that is, when the symbol synchronization timing is shifted by ½symbol, a discontinuous change occurs.

FIG. 63(A) is a schematic view showing an example of the time change ofthe index of the maximum correction value in a case where Fs_t>Fs_r.FIG. 63(B) is a schematic view showing an example of the time change ofthe amplitude of the maximum correction value in the case whereFs_t>Fs_r. FIG. 63(C) is a schematic view showing an example of the timechange of the phase of the maximum correction value in the case whereFs_t>Fs_r.

In the case where Fs_t>Fs_r, since the sample timing of the receivedsignal by the reception apparatus gradually advances with respect to thesample timing of the transmission apparatus, the index decreases as timeelapses. That is, the position of appearance of the maximum correlationvalue is shifted to a sampling position in front.

Moreover, as in the case where Fs_t<Fs_r, the amplitude of the maximumcorrelation value repetitively increases and decreases, and becomesminimum when the index is changed.

Moreover, as in the case where Fs_t<Fs_r, the phase of the maximumcorrelation value is discontinuously changed when the index is changed.In this case, the direction of the phase rotation at the discontinuouspoints is opposite to that in the case where Fs_t<Fs_r, and the phase isrotated by π/2 radians.

Next, the reason why the phase of the maximum correlation value 2110 isrotated when the index of the maximum correlation value 110 is changedwill be described.

FIG. 64 is a schematic view showing an example of the amount of phaserotation by the π/2 shift of the transmission apparatus and the amountof phase reverse rotation by the −π/2 shift of the reception apparatus2000 when the symbol synchronization timings coincide between thetransmission and reception apparatuses.

In FIG. 64, the π/2 shift and the −π/2 shift are canceled out eachother, and the reverse rotation result is 0 radians. The reverserotation result indicates the result of synthesis of the above phaserotation amount and the above phase reverse rotation amount.

FIG. 65 is a schematic view showing an example of the amount of phaserotation by the π/2 shift of the transmission apparatus and the amountof phase reverse rotation by the −π/2 shift of the reception apparatus2000 when the symbol synchronization timing of the reception apparatus2000 is earlier than that of the transmission apparatus by one symbol.

Referring to FIG. 65, it can be understood that since the reverserotation result is −π/2 radians, when the symbol synchronization timingis one symbol earlier, the phase of the maximum correlation value 2110is rotated by −π/2 radians. Here, +3π/2 radians=−π/2 radians.

FIG. 66 is a schematic view showing an example of the amount of phaserotation by the π/2 shift of the transmission apparatus and the amountof phase reverse rotation by the −π/2 shift of the reception apparatus2000 when the symbol synchronization timing of the reception apparatus2000 is later than that of the transmission apparatus by one symbol.

Referring to FIG. 66, it can be understood that since the reverserotation result is +π/2 radians, when the symbol synchronization timingis one symbol later, the phase of the maximum correlation value 2110 isrotated by +π/2 radians. Here, −3π/2 radians=+π/2 radians.

When the symbol synchronization timing is shifted by not less than ½symbol, since the correlation with the adjoining symbol is dominant, thephase of the maximum correlation value 2110 is rotated according to thereverse rotation results of FIG. 65 and FIG. 66.

Moreover, when the symbol synchronization timing of the receptionapparatus 2000 is earlier than that of the transmission apparatus by notless than ½ symbol, the index of the maximum correlation value 2110 isincreased by one (delayed by one sample). On the other hand, when thesymbol synchronization timing of the reception apparatus 2000 is laterthan that of the transmission apparatus by not less than ½ symbol, theindex of the maximum correlation value 2110 is decreased by one(advanced by one sample).

Therefore, it is also said that when the index of the maximumcorrelation value 2110 is increased by one, the phase of the maximumcorrelation value 2110 is rotated by −π/2 radians and when the index ofthe maximum correlation value 2110 is decreased by one, the phase of themaximum correlation value is rotated by +π/2 radians.

Next, using FIG. 67, the correction of the phase of the maximumcorrelation value 2110 by the phase rotator 2107 will be described. FIG.67 illustrates, for simplification of description, a case where there isno carrier frequency error between the transmission and receptionapparatuses. Here, “x” in FIG. 67 indicates that the phase is unknown.

First, attention is focused on a correlation value 2106 _(—) b as themaximum correlation value 2110. The correlation value 2106 _(—) b is areference correlation value here, and is a correlation value obtainedwhen the sample rate is such that Fs_t=Fs_r. It is assumed that theindex of the correlation value 2106 _(—) b is 26 and the phase of thecorrelation value 2106 _(—) b is 0 radians. When the phase of thecorrelation value 2106 _(—) b is rotated by π/2 radians by the phaserotator 2107, the phase 2108 _(—) b of the correlation value 2108 withrespect to the correlation value 2106 _(—) b is π/2 radians.

Since the period of the index is an integral multiple of the phaserotation period of the phase rotator 2107, the phase rotation amount forthe index 26 is always π/2 radians. Likewise, the phase rotation amountfor the index 25 is always 0 radians. Likewise, the phase rotationamount for the index 27 is always π radians. As described above, thephase rotation amounts for the indices are fixed.

Subsequently, attention is focused on a correlation value 2016 _(—) a asthe maximum correlation value 2110. The correlation value 2016 _(—) ais, with the correlation value 2016 _(—) b as the reference, acorrelation value obtained when the sample rate is such that Fs_t>Fs_r.Since the index of the correlation value 2016 _(—) a is lower than theindex of the correlation value 2016 _(—) b by one, the phase is π/2radians. On the other hand, the amount of phase rotation by the phaserotator 2107 is 0 radians, and is smaller than the amount of phaserotation with respect to the correlation value 2106 _(—) b by π/2radians. Therefore, the phase 2108 _(—) a of the correlation value 2108with respect to the correlation value 2016 _(—) a is π/2 radians.

Subsequently, attention is focused on a correlation value 2016 _(—) c asthe maximum correlation value 2110. The correlation value 2016 _(—) cis, with the correlation value 2016 _(—) b as the reference, acorrelation value obtained when the sample rate is such that Fs_t<Fs_r.Since the index of the correlation value 2016 _(—) c is higher by onethan the index of the correlation value 2106 _(—) b, the phase is 3π/2radians. On the other hand, the amount of phase rotation by the phaserotator 2107 is π radians, and is larger than the amount of phaserotation with respect to the correlation value 2016 _(—) b by π/2radians. Therefore, the phase 2108 _(—) c of the correlation value 2108with respect to the correlation value 2016 _(—) c is π/2 radians.

As described above, by shifting the phase of the correlation value 2106by π/2 by the phase rotator 2107, the phase rotation of the maximumcorrelation value 2110 due to the sampling frequency error between thetransmission and reception apparatuses can be corrected.

FIG. 68 is a schematic view showing an example of the result ofcomparison between the phase change of the maximum correlation value2110 and the phase change of the reference phase by the receptionapparatus 2000. The phase of the maximum correlation value 2110 by thereception apparatus 2000 includes the phases of the maximum correlationvalue 2110 in the cases where Fs_t=Fs_r and where Fs_t<Fs_r and in thecase where Fs_t>Fs_r. The reference phase is the phase of the maximumcorrelation value 2110 in the case where Fs_t=Fs_r in the conventionalexample or the present embodiment.

Referring to FIG. 68, it can be understood that according to thereception apparatus 2000, the phase rotation of the maximum correlationvalue 2110 due to the sampling frequency error between the transmissionand reception apparatuses is corrected and substantially coincides withthe reference phase.

The phase rotator 2107 and the maximum value detector 2109 are includedin a maximum correlation value processor 2150. The maximum correlationvalue processor 2150 successively outputs, of the successivelycalculated correlation values, the maximum correlation value which isthe maximum value in the periods divided according to the length of theknown signal stream and is rotated by the rotation amount compliant withthe modulation method.

The carrier frequency error estimator 2111 estimates the carrierfrequency error based on the phase rotation amount of the maximumcorrelation value 2110, and outputs a carrier frequency error estimationvalue 2112. For example, the carrier frequency error estimator 2111calculates a phase rotation amount P between the adjoining two maximumcorrelation values 2110. When the length of the known signal stream isL, the distance between the adjoining maximum correlation values 2110 isL samples, strictly, it is L±1 samples when the index is changed.

For example, the carrier frequency error estimator 2111 divides thephase rotation amount P between the maximum correlation values 2110 by Lto calculate the phase rotation amount per sample. The carrier frequencyerror estimator 2111 outputs the calculation result as the carrierfrequency error estimation value 2112.

The frequency corrector 2113 rotates the phase of the received sample2104 by an amount according to the carrier frequency error estimationvalue 2112, and outputs a received sample 2114 having undergone carrierfrequency error correction.

FIG. 69 is a schematic view showing a first example of the structure ofthe frequency corrector 2113. In FIG. 69, the frequency corrector 2113includes an adder 2113_1, a one sample delay device 2113_3, a signinversion circuit 2113_5, a correction vector generator 2113_7 and acomplex multiplier 2113_9.

The adder 2113_1 and the one sample delay device 2113_3 accumulate thecarrier frequency error estimation value 2112. The sign inversioncircuit 2113_5 inverts the sign of the accumulated carrier frequencyerror estimation value. Thereby, a carrier frequency error correctionvalue 2113_6 for each sample is generated.

The correction vector generator 2113_7 calculates exp(i×θ) and generatesa correction vector 2113_8. Here, “θ is the carrier frequency errorcorrection value 2113_6. The complex multiplier 2113_9 multiplies thereceived sample 2104 and the correction vector 2113_8. Thereby, thephase rotation due to the carrier frequency error is corrected.

FIG. 70 is a schematic view showing a second example of the structure ofthe frequency corrector 2113. FIG. 70 is a structure where thecorrection vector generator 2113_7 and the complex multiplier 2113_9 ofFIG. 69 are replaced by a CORDIC calculation circuit 2113_11. Since theCORDIC algorithm uses no multiplier, in the structure of FIG. 70, thecircuit size can be reduced compared with the structure of FIG. 69, sothat power consumption can be reduced.

According to the reception apparatus 2000, the phase rotation of themaximum correlation value 2110 due to the shift of the phase reverserotation timing (sampling frequency error) of the phase reverse rotator2103 can be corrected by using the maximum correlation value rotated bythe rotation amount compliant with the modulation method by the maximumcorrelation value processor 2150. For example, by the phase rotator 2107rotating the phase of the correlation value 2106 by the rotation amountcompliant with the modulation method, the phase rotation of the maximumcorrelation value 2110 due to the sampling frequency error can becorrected. Therefore, even when a sampling frequency error is presentbetween the transmission and reception apparatuses, the precision of theestimation of the carrier frequency error between the transmission andreception apparatuses can be improved, and the precision of the carrierfrequency error correction can be improved. Moreover, even inasynchronous condition, the phase of the correlation peak can bestabilized.

The sampling section 2101 may perform N-times (N is an integer)oversampling of the received signal 2100 (that is, Fs_r≅N*Rsym). In thiscase, the phase reverse rotator 2103 shifts the received sample 2102 by−π/2N, that is, rotates it by −π/2N radians. The phase rotator 2107shifts the correlation value 2106 by π/2N, that is, rotates it by π/2N.The carrier frequency error estimator 2111 divides, for example, thephase rotation amount P between the maximum correlation values by N*L.Thereby, even when N-times oversampling is performed, the precision ofthe estimation of the carrier frequency error between the transmissionand reception apparatuses can be improved, and the precision of thecarrier frequency error correction can be improved.

When a modulation method where the phase is rotated by π/M (M is aninteger) is used, the phase reverse rotator 2103 shifts the receivedsample 2102 by −π/M, and the phase rotator 2107 shifts the correlationvalue 2106 by π/M. Thereby, even when the modulation method where thephase is rotated by π/M (M is an integer) is used, the precision of theestimation of the carrier frequency error between the transmission andreception apparatuses can be improved, and the precision of the carrierfrequency error correction can be improved.

Thirteenth Embodiment

FIG. 71 is a block diagram showing a structure example of a receptionapparatus 2000B in a thirteenth embodiment.

The reception apparatus 2000B is provided with the sampling section2101, the phase reverse rotator 2103, the correlation value calculator2105, a maximum value detector 2115, a phase rotator 2117, the carrierfrequency error estimator 2111 and the frequency corrector 2113.

A difference between FIG. 56 and FIG. 71 is that the positions of thephase rotator and the maximum value detector are reversed. In thereception apparatus 2000B of FIG. 71, elements the same as those of thereception apparatus 2000 of FIG. 56 are denoted by the same referencenumerals and signs and descriptions thereof are omitted or simplified.

The maximum value detector 2115 and the phase rotator 2117 are includedin a maximum correlation value processor 2160 as a modification of themaximum correlation value processor 2150.

Like the maximum value detector 2109 of the twelfth embodiment, when thelength of the known signal stream is L, the maximum value detector 2115detects a maximum correlation value 2116 _(—) a in the L sample periodfrom among the successively derived correlation values 2106. The maximumvalue detector 2115 outputs the maximum correlation value 2116 _(—) aand the index 2116 _(—) b of the maximum correlation value.

The phase rotator 2117 rotates the phase of the maximum correlationvalue 2116 _(—) a by the rotation amount compliant with the modulationmethod, and outputs a phase-rotated maximum correlation value 2118.

Moreover, the phase rotator 2117 may control the phase rotation amountby which the phase of the maximum correlation value 2116 _(—) a isrotated, according to the change of the index 2116 _(—) b of the maximumcorrelation value. Thereby, the phase rotation by the phase rotator 2117can be sometimes omitted, so that the processing load on the phaserotator 2117 can be reduced.

FIG. 72 is a block diagram showing a structure example of the phaserotator 2117. The phase rotator 2117 includes a delay section 2119, acomparator 2121, a rotation amount controller 2123 and a rotator 2125.

The delay section 2119 delays the index 2106 _(—) b of the maximumcorrelation value by one sample, and outputs the index 2120 of themaximum value time-delayed by one sample. The comparator 2121 comparesthe current index with the one sample preceding index, and outputs thecomparison result 2122.

For example, when the current index and the one sample preceding indexare the same, “0” is outputted as the comparison result 2122. When thecurrent index is higher than the one sample preceding index, “+1” isoutputted as the comparison result 2122. When the current index is lowerthan the one sample preceding index, “−1” is outputted as the comparisonresult 2122.

The rotation amount controller 2123 controls a phase rotation amount2124 according to the comparison result 2122. For example, when thecomparison result 2122 is “0”, the phase rotation amount 2124 is held.That is, when the current index and the one sample preceding index arethe same, the rotation amount controller 2123 does not change thecurrent phase rotation amount.

Moreover, when the comparison result 2122 is “+1”, the rotation amountcontroller 2123 adds +π/2 radians to the phase rotation amount 2124.That is, when the current index is higher than the one sample precedingindex, the rotation amount controller 2123 increases the phase rotationamount.

Moreover, when the comparison result 2122 is “−1”, the rotation amountcontroller 2123 adds −π/2 radians to the phase rotation amount 2124.That is, when the current index is lower than the one sample precedingindex, the rotation amount controller 2123 reduces the phase rotationamount.

The rotator 2125 rotates the phase of the maximum correlation value 2116_(—) a by the phase rotation amount 2124, and outputs the phase-rotatedmaximum correlation value 2118.

According to the reception apparatus 2000B, by rotating the maximumcorrelation value 2116 _(—) a by the rotation amount compliant with themodulation method after detecting the maximum correlation value 2116_(—) a, the phase rotation of the maximum correlation value 2116 _(—) adue to the sampling frequency error can be corrected. For example, thephase of the maximum correlation value 2116 _(—) a is rotated by usingthe phase rotation amount 2124 controlled according to the change of theindex 2116 _(—) b of the maximum correlation value detected by themaximum value detector 2115. Thereby, the phase rotation of the maximumcorrelation value 2116 _(—) a due to the shift of the phase reverserotation timing (sampling frequency error) of the phase reverse rotator2103 can be corrected. Therefore, even when a sampling frequency erroris present between the transmission and reception apparatuses, theprecision of the estimation of the carrier frequency error between thetransmission and reception apparatuses can be improved, and theprecision of the carrier frequency error correction can be improved.

The sampling section 2101 may perform N-times (N is an integer)oversampling of the received signal 2100 (that is, Fs_r≅N*Rsym). In thiscase, when “the comparison result 2122=+1”, the rotation amountcontroller 2123 adds +π/2N radians to the phase rotation amount 2124,and when “the comparison result 2122=−1”, the rotation amount controller2123 adds −π/2N radians to the phase rotation amount 2124.

When the modulation method where the phase is rotated by π/M (M is aninteger) is used, when “the comparison result 2122=+1”, the rotationamount controller 2123 adds +π/M radians to the phase rotation amount2124, and when “the comparison result 2122=−1”, the rotation amountcontroller 2123 adds −π/M radians to the phase rotation amount 2124.

While various embodiments have been described with reference to thedrawings, it is to be noted that the present disclosure is not limitedto such examples. It is obvious that one of ordinary skill in the artcan arrive at various change examples or modification examples withinthe category described in the claims, and it is to be understood thatthose naturally belong to the technical scope of the present disclosure.Moreover, elements in the above-described embodiments may be arbitrarilycombined without departing from the gist of the disclosure.

While in the above-described embodiments, the present disclosure isdescribed by using as an example a case of implementation usinghardware, the present disclosure may also be implemented by software inconjunction with hardware.

Moreover, the functional blocks used for the description of theembodiments are typically implemented as an LSI which is an integratedcircuit. These may be individually formed as one chip, or may be formedas one chip so as to include some or all of the functional blocks. WhileLSI is mentioned here, it is sometimes called IC, system LSI, super LSIor ultra LSI according to the difference in integration degree.

Moreover, the method of circuit integration is not limited to LSI but adedicated circuit or a general-purpose processor may be used forimplementation. An FPGA (Field Programmable Gate Array) whereprogramming can be performed after LSI manufacture or a reconfigurableprocessor where the connection and setting of the circuit cells in theLSI are reconfigurable may be used.

Further, if a circuit integration technique replacing the LSI comes outwith the advancement of the semiconductor technology or by a differenttechnology being derived, as a natural consequence, the functionalblocks may be integrated by using the different technique. Theapplication of biotechnology or the like has a potential.

The present disclosure may be expressed as a phase error estimationmethod or a phase error correction method executed in a receptionapparatus or a wireless communication apparatus. Moreover, the presentdisclosure may be expressed as a phase error estimation apparatus or aphase error correction apparatus as an apparatus that has the functionto execute a phase error estimation method or a phase error correctionmethod, or a program that causes a computer to operate a phase errorestimation method or apparatus or a phase error correction method orapparatus. That is, the present disclosure may be expressed in any ofthe apparatus, method and program categories.

SUMMARY OF ONE MODE OF THE PRESENT DISCLOSURE

(1) A reception apparatus having:

a phase error estimator that extracts a specific reference signal from areceived signal which is a received transmitted signal having thespecific reference signal, and obtains a received reference signal in afrequency domain in a receiver,

-   -   compares, for each frequency, the received reference signal in        the frequency domain and a transmitted reference signal which is        the specific reference signal in the transmitted signal        expressed in the frequency domain, and obtains a plurality of        error vectors, and

obtains, from the error vectors, a phase error inclination and a phaseerror offset in the frequency domain possessed by the received referencesignal, and estimates a phase error according to the frequency by thephase error inclination and the phase error offset; and

a phase error corrector that corrects the phase error for the receivedsignal by using a phase error estimation value obtained by the phaseerror estimator.

(2) A phase error estimation method, having:

extracting a specific reference signal from a received signal which is areceived transmitted signal having the specific reference signal, andobtaining a received reference signal in a frequency domain in areceiver;

comparing, for each frequency, the received reference signal in thefrequency domain and a transmitted reference signal which is thespecific reference signal in the transmitted signal expressed in thefrequency domain, and obtaining a plurality of error vectors;

dividing the error vectors into not less than two groups, obtaining arepresentative value of each group and obtaining a plurality ofrepresentative vectors; and

obtaining the phase error inclination and the phase error offset in thefrequency domain possessed by the received reference signal based on therepresentative vectors, and estimating a phase error according to thefrequency by the phase error inclination and the phase error offset.

(3) The above-described phase error estimation method, wherein in theestimation of the phase error, the representative vectors are convertedfrom vectors into phases, and when the phases between the representativevectors are discontinuous, a predetermined phase value is added orsubtracted when the phase error inclination and the phase error offsetare obtained.

(4) The above-described phase error estimation method, wherein in theobtaining of the representative vectors, the groups are dividedaccording to a magnitude of the frequency with respect to the errorvectors, and are groups of a predetermined number of error vectorsextracted in decreasing order of a magnitude of an amplitude.

(5) The above-described phase error estimation method, wherein in theobtaining of the representative vectors, as the representative value ofeach group, a mean value is obtained by vector mean of the errorvectors.

(6) The above-described phase error estimation method, wherein in theobtaining of the representative vectors, the mean value is obtained byadding the error vectors.

(7) The above-described phase error estimation method, wherein in theobtaining of the representative vectors, by using a plurality of errorvectors to which a predetermined weight is assigned according to amagnitude of an amplitude by the frequency with respect to the errorvectors, the mean value is obtained by adding the error vectors.

(8) The above-described phase error estimation method, wherein in theestimation of the phase error, it is determined that the phases betweenthe representative vectors are discontinuous when an absolute value of aphase difference between two target representative vectors of therepresentative vectors is not less than π.

(9) The above-described phase error estimation method, wherein in theestimation of the phase error, the representative vectors are convertedfrom vectors into phases, and for phases of two target representativevectors of the representative vectors, wrapping subtraction of the phaseof a low frequency from the phase of a high frequency is performed sothat a calculation result falls within a range of ±π and division by afrequency difference between the two representative vectors is performedto thereby obtain the phase error inclination.

(10) The above-described phase error estimation method, wherein in theestimation of the phase error, the representative vectors are convertedfrom vectors into phases, phases of two target representative vectors ofthe representative vectors are added and gain multiplication by ½ isperformed, and when the phases between the representative vectors arediscontinuous, wrapping addition of π to a result of the gainmultiplication is performed so that a calculation result falls within arange of ±π to thereby obtain the phase error offset.

(11) The above-described phase error estimation method,

wherein in the obtaining of the representative vectors, the groups aredivided into not less than three groups and a representative value ofeach group is obtained, and

in the estimation of the phase error, the representative vectors areconverted from vectors into phases, unwrapping processing is performedon the phases after the conversion and linear approximation by an LSM isperformed to thereby obtain the phase error inclination and offset.

(12) A reception method having: obtaining a phase error according to thefrequency by the phase error estimation of the phase error estimationmethod described in any of the above; and performing correction of thephase error on the received signal.

(13) A reception method having: obtaining a phase error according to thefrequency by the phase error estimation of the phase error estimationmethod described in any of the above;

obtaining a phase by a transmission characteristic possessed by atransmission path between a transmitter and a receiver by transmissionpath estimation; and

performing phase correction of the phase error obtained by the phaseerror estimation and the phase of the transmission characteristicobtained by the transmission path estimation together.

(14) A phase error estimation apparatus having:

a signal extractor that extracts a specific reference signal from areceived signal which is a received transmitted signal having thespecific reference signal, and obtains a received reference signal in afrequency domain in a receiver;

an error vector calculator that compares, for each frequency, thereceived reference signal in the frequency domain and a transmittedreference signal which is the specific reference signal in thetransmitted signal expressed in the frequency domain, and obtains aplurality of error vectors;

a representative vector calculator that divides the error vectors intonot less than two groups, obtains a representative value of each groupand obtains a plurality of representative vectors; and

a correction value calculator that obtains the phase error inclinationand the phase error offset in the frequency domain possessed by thereceived reference signal based on the representative vectors, andestimates a phase error according to the frequency by the phase errorinclination and the phase error offset.

(15) A reception apparatus having:

the above-described phase error estimation apparatus; and

a phase corrector that performs correction of the phase error on thereceived signal.

(16) A reception apparatus having:

the above-described phase error estimation apparatus;

a transmission path estimator that obtains a phase by a transmissioncharacteristic possessed by a transmission path between a transmitterand a receiver by transmission path estimation; and

a transmission path corrector that performs phase correction of thephase error obtained by the phase error estimation apparatus and thephase of the transmission characteristic obtained by the transmissionpath estimator together.

(17) A carrier frequency offset correction method having: extracting aspecific reference signal from a received signal which is a receivedtransmitted signal having the specific reference signal, and obtaining areceived reference signal in a frequency domain after rough carrierfrequency offset correction and rough symbol synchronization shiftcorrection in a receiver;

comparing, for each frequency, the received reference signal in thefrequency domain and a transmitted reference signal which is thespecific reference signal in the transmitted signal expressed in thefrequency domain, and obtaining a plurality of error vectors;

estimating a phase error by linear approximation in the frequency domainfrom the error vectors, and obtaining the phase error inclination as aresidual symbol synchronization shift and the phase error offset as aresidual carrier frequency offset;

performing correction of the residual symbol synchronization shift inthe frequency domain; and

performing correction of the residual carrier frequency offset on thereceived signal converted from the frequency domain into the timedomain.

(18) A carrier frequency offset correction method having:

extracting a specific reference signal from a received signal which is areceived transmitted signal having the specific reference signal, andobtaining a received reference signal in a frequency domain after roughcarrier frequency offset correction and rough symbol synchronizationshift correction in a receiver;

comparing, for each frequency, the received reference signal in thefrequency domain and a transmitted reference signal which is thespecific reference signal in the transmitted signal expressed in thefrequency domain, and obtaining a plurality of error vectors;

estimating a phase error by linear approximation in the frequency domainfrom the error vectors, and obtaining the phase error inclination as aresidual symbol synchronization shift and the phase error offset as aresidual carrier frequency offset; and

performing correction of the residual symbol synchronization shift andcorrection of the residual carrier frequency offset on the receivedsignal converted from the frequency domain into the time domain.

(19) The above-described carrier frequency offset correction method,wherein in the estimation of the phase error, the error vectors areconverted from vectors into phases, unwrapping processing is performedon the phases after the conversion and linear approximation isperformed.

(20) The above-described carrier frequency offset correction method,wherein in the correction of the residual carrier frequency offset inthe time domain, the phase error offset is converted from a phase into avector, a conjugate complex number of the vector after the conversion isset as a correction value and the correction value and the receivedsignal in the time domain are complex-multiplied to thereby correct theresidual carrier frequency offset.

(21) The above-described carrier frequency offset correction method,wherein in the correction of the residual carrier frequency offset inthe time domain, a phase which is the phase error offset multiplied by−1 is set as a correction value and a CORDIC calculation of thecorrection value and the received signal in the time domain together isperformed to thereby correct the residual carrier frequency offset.

(22) The above-described carrier frequency offset correction method,wherein in the correction of the residual symbol synchronization shiftin the frequency domain, the phase error inclination and a coefficientcorresponding to the frequency are multiplied to obtain a symbolsynchronization shift amount of each frequency and for each frequency,the symbol synchronization shift amount is converted from a phase into avector, a conjugate complex number of the vector after the conversion isset as a correction value and the correction value and the receivedsignal in the frequency domain are complex-multiplied to thereby correctthe residual symbol synchronization shift.

(23) The above-described carrier frequency offset correction method,wherein in the correction of the residual symbol synchronization shiftin the frequency domain, the phase error inclination and a coefficientcorresponding to the frequency are multiplied to obtain a symbolsynchronization shift amount of each frequency and for each frequency, aphase which is the symbol synchronization shift amount multiplied by −1is set as a correction value and a CORDIC calculation of the correctionvalue and the received signal in the frequency domain together isperformed to thereby correct the residual symbol synchronization shift.

(24) The above-described carrier frequency offset correction method,wherein in the correction of the residual symbol synchronization shiftin the time domain, a filter performing synchronization correction isused, the phase error inclination is converted into a symbolsynchronization shift amount, a filter coefficient corresponding to thesymbol synchronization shift amount is selected and the received signalin the time domain is passed through the filter to thereby correct theresidual symbol synchronization shift.

(25) The above-described carrier frequency offset correction method,wherein in the time domain, after the correction of the residual carrierfrequency offset is performed, the correction of the residual symbolsynchronization shift is performed.

(26) The above-described carrier frequency offset correction method,wherein in the time domain, after the correction of the residual symbolsynchronization shift is performed, the correction of the residualcarrier frequency offset is performed.

(27) A carrier frequency offset correction apparatus having:

a signal extractor that extracts a specific reference signal from areceived signal which is a received transmitted signal having thespecific reference signal, and obtains a received reference signal in afrequency domain after rough carrier frequency offset correction andrough symbol synchronization shift correction in a receiver;

an error vector calculator that compares, for each frequency, thereceived reference signal in the frequency domain and a transmittedreference signal which is the specific reference signal in thetransmitted signal expressed in the frequency domain, and obtains aplurality of error vectors;

a residual phase error calculator that estimates a phase error by linearapproximation in the frequency domain from the error vectors, andobtains the phase error inclination as a residual symbol synchronizationshift and the phase error offset as a residual carrier frequency offset;

a residual symbol synchronization shift corrector that performscorrection of the residual symbol synchronization shift in the frequencydomain;

a frequency-time converter that converts the received signal in thefrequency domain into a time domain; and

a residual carrier frequency offset corrector that performs correctionof the residual carrier frequency offset on the received signalconverted from the frequency domain into the time domain.

(28) A carrier frequency offset correction apparatus having:

a signal extractor that extracts a specific reference signal from areceived signal which is a received transmitted signal having thespecific reference signal, and obtains a received reference signal in afrequency domain after rough carrier frequency offset correction andrough symbol synchronization shift correction in a receiver;

an error vector calculator that compares, for each frequency, thereceived reference signal in the frequency domain and a transmittedreference signal which is the specific reference signal in thetransmitted signal expressed in the frequency domain, and obtains aplurality of error vectors;

a residual phase error calculator that estimates a phase error by linearapproximation in the frequency domain from the error vectors, andobtains the phase error inclination as a residual symbol synchronizationshift and the phase error offset as a residual carrier frequency offset;

a frequency-time converter that converts the received signal in thefrequency domain into a time domain;

a residual carrier frequency offset corrector that performs correctionof the residual carrier frequency offset on the received signalconverted from the frequency domain into the time domain; and

a time domain residual symbol synchronization shift corrector thatperforms correction of the residual symbol synchronization shift on thereceived signal converted from the frequency domain into the timedomain.

(29) A reception apparatus that receives a signal modulated by amodulation method where a phase is rotated, from a transmissionapparatus, the reception apparatus having:

a phase reverse rotator that reversely rotates a phase of a signal whichis a received signal sampled at a predetermined sampling frequency, by arotation amount according to the modulation method;

a correlator that successively calculates a correlation value between asignal whose phase is reversely rotated by the phase reverse rotator anda predetermined signal;

a maximum correlation value processor that successively outputs amaximum correlation value which is, of the correlation valuessuccessively calculated by the correlator, a maximum value in a perioddivided according to a length of the predetermined signal, and that isrotated by a rotation amount according to the modulation method;

a carrier frequency error estimator that estimates a carrier frequencyerror between the transmission apparatus and the reception apparatusaccording to an amount of phase rotation between a plurality of maximumcorrelation values successively outputted from the maximum correlationvalue processor; and

a carrier frequency error corrector that rotates the phase of thesampled signal according to the carrier frequency error estimated by thecarrier frequency error estimator.

(30) The above-described receiver,

wherein the maximum correlation value processor is provided with:

a phase rotator that rotates a phase of the correlation valuesuccessively calculated by the correlator, by the rotation amountaccording to the modulation method; and

a maximum value detector that detects, of the correlation values whosephases are rotated by the phase rotator, the maximum value included inthe period, as the maximum correlation value.

(31) The above-described reception apparatus,

wherein the maximum correlation value processor is provided with:

a maximum value detector that detects, of the correlation valuessuccessively calculated by the correlator, the maximum value included inthe period according to the length of the predetermined signal, as themaximum correlation value; and

a phase rotator that rotates a phase of the maximum correlation valuedetected by the maximum value detector, by the rotation amount accordingto the modulation method.

(32) The above-described reception apparatus, wherein the phase rotatorcontrols the phase rotation amount by which the phase of the maximumcorrelation value is rotated, according to a change of an indexrepresentative of a position of the maximum correlation value in theperiod.

(33) The above-described reception apparatus, wherein the modulationmethod includes π/2 shift BPSK (Binary Phase Shift Keying).

(34) The above-described reception apparatus,

wherein the phase reverse rotator reversely rotates the phase of anN-times oversampled received signal, by −π/2 radians, and

the phase rotator rotates the phase of the correlation value by π/2Nradians.

(35) A reception method in a reception apparatus that receives a signalmodulated by a predetermined modulation method, from a transmissionapparatus, the reception method having:

a step of reversely rotating a phase of a signal which is a receivedsignal sampled at a predetermined sampling frequency, by a rotationamount according to the modulation method;

a step of successively calculating a correlation value between a signalwhose phase is reversely rotated and a predetermined signal;

a step of successively outputting a maximum correlation value that isincluded in the period according to the length of the predeterminedsignal, is a maximum value of the successively calculated correlationvalues, and is rotated by a rotation amount according to the modulationmethod;

a step of estimating a carrier frequency error between the transmissionapparatus and the reception apparatus according to an amount of phaserotation between the successively outputted maximum correlation values;and

a step of rotating the phase of the sampled signal according to theestimated carrier frequency error.

The present application is based upon Japanese Patent Application (No.2013-037684) filed on Feb. 27, 2013, Japanese Patent Application (No.2013-041054) filed on Mar. 1, 2013 and Japanese Patent Application (No.2013-049364) filed on Mar. 12, 2013, the contents of which areincorporated herein by reference.

INDUSTRIAL APPLICABILITY

The present disclosure has an advantage in that a highly precise phasecorrection value can be obtained even if the phase error or the noiselevel is high. Moreover, the present disclosure has an advantage in thatthe carrier frequency offset can be appropriately corrected and thereceived data can be correctly demodulated. The present disclosure isuseful, for example, as a wireless apparatus performing high-speedtransmission, a phase error estimation method and a phase errorcorrection method applied to a wireless apparatus.

DESCRIPTION OF REFERENCE NUMERALS AND SIGNS

1 RF processor

2 ADC section

4 Synchronization detector

5 Frequency corrector

6 S/P converter

7 DFT section

8, 8A Transmission path corrector

80 Transmission path estimator

81-00 to 81-63 Multiplier

82-00 to 82-63 Rotator

83-00 to 83-63 Wrapping adder

84-00 to 84-63 Flip-flop circuit

85-00 to 85-63 Wrapping adder

9 Phase error corrector

90 Signal extractor

91 Error vector calculator

910-00 to 910-07 Complex multiplier

92 Representative vector calculator

920-L, 920-H Complex adder

93 Correction value calculator

930-L, 930-H, 930-M Vector-phase converter

931, 931-LM, 931-MH Phase inclination calculator

9310 Wrapping subtractor

9311 Gain multiplicator

932, 932-LM, 932-MH Phase offset calculator

9320 Adder

9321 Gain multiplier

9322 Wrapping adder

9323 Selector

9324 Determiner

93240 Adder

93241 a, 93241 b Inequality sign determiner

93242 OR circuit

933 Frequency-by-frequency correction value calculator

9330-00 to 9330-63 Multipliers

9331-00 to 9331-63 Adders

934 Phase inclination averaging section

9340 Adder

9341 Gain multiplicator

935 Phase offset averaging section

9350 Adder

9351 Gain multiplicator

9352 Wrapping adder

9353 Selector

9354 Determiner

938 Phase unwrapping section

939 LSM approximator

94 Phase corrector

940-00 to 940-63 Phase-vector converter

941-00 to 941-63 Conjugate converter

942-00 to 942-63 Complex multiplier

95 Phase error estimator

10 IDFT section

11 P/S converter

13 Demodulator

15 Selector

1001 RF processor

1002 ADC section

1003 AGC section

1004 Synchronization detector

1005 Frequency corrector

1006 S/P converter

1007 DFT section

1008 Transmission path corrector

1009 Phase error corrector

1090 Signal extractor

1091 Error vector calculator

1910-00 to 1910-07 Complex multiplier

1092 Phase error calculator

1920-00 to 1920-07 Vector-phase converter

1021-00 to 1921-07 Unwrapping section

1093 Residual phase error calculator

1094 Residual symbol synchronization shift calculator

1940-00 to 1940-63 Multiplier

1095 Residual symbol synchronization shift corrector

1950-00 to 1950-63 Phase-vector converter

1951-00 to 1951-63 Conjugate converter

1952-00 to 1952-63 Complex multiplier

1953-00 to 1953-63 Gain multiplicator

1954-00 to 1954-63 CORDIC section

1010 IDFT section

1011 P/S converter

1012, 1012A Residual carrier frequency offset corrector

1120 Phase-vector converter

1121 Conjugate converter

1122 Complex multiplier

1123 Gain multiplicator

1124 CORDIC section

1013 Demodulator

1015 Phase error estimator

1016 Time domain residual symbol synchronization shift corrector

1160 IQ separator

1161 Correction coefficient selector

1162-I-00 to 1162-I-09, 1162-Q-00 to 1162-Q-09 FF section

1163-1-00 to 1163-I-10, 1163-Q-00 to 1163-Q-10 Multiplier

1164-I, 1164-Q Adder

1165 IQ unification section

2000, 2000B Reception apparatus

2100 Received signal

2101 Sampling section

2102 Received sample

2103 Phase reverse rotator

2104 Phase-reversely-rotated received sample

2105 Correlation value calculator

2106 Correlation value

2106 _(—) i I component of correlation value 2016

2016 _(—) q Q component of correlation value 2016

2016 _(—) a Correlation value

2016 _(—) b Correlation value

2016 _(—) c Correlation value

2107 Phase rotator

2107_1 a Cosine wave generator

2107_1 b Sinusoidal wave generator

2107_3 a Multiplier

2107_3 b Multiplier

2107_3 c Multiplier

2107_3 d Multiplier

2107_5 a Multiplier

2107_5 b Multiplier

2107_7 Counter

2107_9 Selector

2107_10 Phase information

2107_11 CORDIC calculation circuit

2108 Phase-rotated correlation value

2108 _(—) i I component of correlation value 2108

2108 _(—) q Q component of correlation value 2108

2108 _(—) a Phase of correlation value 2108

2108 _(—) b Phase of correlation value 2108

2108 _(—) c Phase of correlation value 2108

2109 Maximum value detector

2110 Maximum correlation value

2111 Carrier frequency error estimator

2112 Carrier frequency error estimation value

2113 Frequency corrector

2113_1 Adder

2113_3 One sample delay device

2113_5 Sign inversion circuit

2113_6 Carrier frequency error correction value

2113_7 Correction vector generator

2113_8 Correction vector

2113_9 Complex multiplier

2113_11 CORDIC calculation circuit

2114 Received sample having undergone carrier frequency error correction

2115 Maximum value detector

2116 _(—) a Maximum correlation value

2116 _(—) b Index of maximum correlation value

2117 Phase rotator

2118 Phase-rotated maximum correlation value

2119 Delay section

2120 Index of maximum value time-delayed by one sample

2121 Comparator

2122 Comparison result

2123 Rotation amount controller

2124 Phase rotation amount

2125 Rotator

2150 Maximum correlation value processor

1. A reception apparatus comprising: a phase error estimator thatestimates a phase error estimation value in a frequency domain; and aphase error corrector that corrects the phase error for the receivedsignal by using the phase error estimation value, wherein the phaseerror estimator includes that a signal extractor extracts a receivedreference signal in the frequency domain from a received signalincluding a specific reference signal; an error vector calculatorobtains a plurality of error vectors by comparing the received referencesignal in the frequency domain and a transmitted reference signal in thefrequency domain corresponding to the specific reference signal in atransmitter; and a correction value calculator obtains, based on theplurality of error vectors, an amount of phase error change and a phaseerror offset in the frequency domain possessed by the received referencesignal and obtains the phase error estimation value in the frequencydomain based on the amount of phase error change and the phase erroroffset.
 2. The reception apparatus according to claim 1, wherein thephase error estimator further includes that a representative vectorcalculator divides the error vectors into not less than two groups,obtains a plurality of representative vectors as a representative valueof each group, and the correction value calculator obtains the amount ofphase error change and the phase error offset based on the plurality ofrepresentative vectors.
 3. A phase error estimation method of areception apparatus, comprising: extracting a received reference signalin a frequency domain in a receiver from a received signal including aspecific reference signal; obtaining a plurality of error vectors bycomparing the received reference signal in the frequency domain and atransmitted reference signal in the frequency domain corresponding tothe specific reference signal in a transmitter; dividing the errorvectors into not less than two groups, a plurality of representativevectors as a representative value of each group; and obtaining an amountof phase error change and a phase error offset in the frequency domainpossessed by the received reference signal based on the plurality ofrepresentative vectors, and obtaining a phase error value in thefrequency domain based on the amount of phase error change and the phaseerror offset.
 4. The phase error estimation method according to claim 3,wherein in the obtaining of the phase error value, the representativevectors are converted from vectors into phases, and when the phasesbetween the representative vectors are discontinuous, a predeterminedphase value is added or subtracted to the phases.
 5. The phase errorestimation method according to claim 3, wherein in the obtaining of therepresentative vectors, the groups are divided according to a magnitudeof a frequency of each error vector, and are groups of a predeterminednumber of error vectors extracted in decreasing order of a magnitude ofan amplitude of representative vector.
 6. The phase error estimationmethod according to claim 3, wherein in the obtaining of therepresentative vectors, as the representative value of each group, amean value is obtained by vector mean of the error vectors.
 7. The phaseerror estimation method according to claim 6, wherein in the obtainingof the representative vectors, the mean value is obtained by adding theerror vectors.
 8. The phase error estimation method according to claim6, wherein in the obtaining of the representative vectors, by using aplurality of error vectors to which a predetermined weight is assignedaccording to a magnitude of an amplitude of each error vector in thefrequency domain, the mean value is obtained by adding the errorvectors.
 9. The phase error estimation method according to claim 4,wherein in the obtaining of the phase error, it is determined that thephases between the representative vectors are discontinuous when anabsolute value of a phase difference between two target representativevectors of the representative vectors is not less than π.
 10. The phaseerror estimation method according to claim 3, wherein in the obtainingof the phase error, the representative vectors are converted fromvectors into phases, and for phases of two target representative vectorsof the representative vectors, wrapping subtraction of the phase of alow frequency from the phase of a high frequency is performed so that acalculation result falls within a range of ±π and division by afrequency difference between the two representative vectors isperformed.
 11. The phase error estimation method according to claim 3,wherein in the estimation of the phase error, the representative vectorsare converted from vectors into phases, phases of two targetrepresentative vectors of the representative vectors are added and gainmultiplication by ½ is performed, and when the phases between therepresentative vectors are discontinuous, wrapping addition of π to aresult of the gain multiplication is performed so that a calculationresult falls within a range of ±π.
 12. The phase error estimation methodaccording to claim 3, wherein in the obtaining of the representativevectors, the groups are divided into not less than three groups and arepresentative value of each group is obtained; and wherein in theestimation of the phase error, the representative vectors are convertedfrom vectors into phases, unwrapping processing is performed on thephases after the conversion and linear approximation by an LSM isperformed.
 13. The reception apparatus according to claim 1, wherein thesignal extractor extracts the received reference signal after roughcarrier frequency offset correction and rough symbol synchronizationshift correction; wherein the correction value calculator obtains thephase error estimation value with linear approximation by obtaining theamount of phase error change as a residual symbol synchronization shiftand the phase error offset as a residual carrier frequency offset; andwherein the phase error corrector performs correction of the residualsymbol synchronization shift in the frequency domain, converts thereceived signal in the frequency domain to a time domain, and performscorrection of the residual carrier frequency offset on the receivedsignal converted from the frequency domain to the time domain.
 14. Thereception apparatus according to claim 1, wherein the signal extractorextracts the received reference signal after rough carrier frequencyoffset correction and rough symbol synchronization shift correction;wherein the correction value calculator obtains the phase errorestimation value with linear approximation by obtaining the amount ofphase error change as a residual symbol synchronization shift and thephase error offset as a residual carrier frequency offset; and whereinthe phase error corrector converts the received signal in the frequencydomain into a time domain, performs correction of the residual carrierfrequency offset on the received signal converted from the frequencydomain into the time domain, and performs correction of the residualsymbol synchronization shift on the received signal converted from thefrequency domain into the time domain.
 15. A phase error correctionmethod of a reception apparatus, comprising: extracting a receivedreference signal in a frequency domain from a received signal includinga specific reference signal after rough carrier frequency offsetcorrection and rough symbol synchronization shift correction in areceiver; obtaining a plurality of error vectors by comparing thereceived reference signal in the frequency domain and a transmittedreference signal in the frequency domain corresponding to the specificreference signal in a transmitter; estimating a phase error estimationvalue with linear approximation by obtaining an amount of phase errorchange as a residual symbol synchronization shift and the phase erroroffset as a residual carrier frequency offset; performing correction ofthe residual symbol synchronization shift in the frequency domain; andperforming correction of the residual carrier frequency offset on thereceived signal converted from the frequency domain into the timedomain.
 16. A phase error correction method of a reception apparatus,comprising: extracting a received reference signal in a frequency domainfrom a received signal including a specific reference signal after roughcarrier frequency offset correction and rough symbol synchronizationshift correction in a receiver; obtaining a plurality of error vectorsby comparing the received reference signal in the frequency domain and atransmitted reference signal in the frequency domain corresponding tothe specific reference signal in a transmitter; estimating a phase errorestimation value with linear approximation by obtaining an amount ofphase error change as a residual symbol synchronization shift and thephase error offset as a residual carrier frequency offset; andperforming correction of the residual symbol synchronization shift andcorrection of the residual carrier frequency offset on the receivedsignal converted from the frequency domain into the time domain.
 17. Thephase error correction method according to claim 15, wherein in theestimating of the phase error estimation value, the plurality of errorvectors are converted from vectors into phases, unwrapping processing isperformed on the phases after the conversion and linear approximation isperformed.
 18. The phase error correction method according to claim 15,wherein in the performing correction of the residual carrier frequencyoffset in the time domain, the phase error offset is converted from aphase into a vector, a conjugate complex number of the vector after theconversion is set as a correction value and the correction value and thereceived signal in the time domain are complex-multiplied.
 19. The phaseerror correction method according to claim 15, wherein in the performingcorrection of the residual carrier frequency offset in the time domain,a phase which is the phase error offset multiplied by −1 is set as acorrection value and a CORDIC calculation of the correction value andthe received signal in the time domain together is performed.
 20. Thephase error correction method according to claim 15, wherein in theperforming correction of the residual symbol synchronization shift inthe frequency domain, the amount of phase error change and a coefficientin the frequency domain are multiplied to obtain an amount of theresidual symbol synchronization shift, the amount of the residual symbolsynchronization shift is converted from a phase into a vector, aconjugate complex number of the vector after the conversion is set as acorrection value and the correction value and the received signal in thefrequency domain are complex-multiplied.
 21. The phase error correctionmethod according to claim 15, wherein in the performing correction ofthe residual symbol synchronization shift in the frequency domain, theamount of phase error change and a coefficient corresponding to thefrequency of each error vector are multiplied to obtain an amount of theresidual symbol synchronization shift, a phase which is the amount ofthe symbol synchronization shift multiplied by −1 is set as a correctionvalue and a CORDIC calculation of the correction value and the receivedsignal in the frequency domain together is performed.
 22. The phaseerror correction method according to claim 16, wherein in the performingcorrection of the residual symbol synchronization shift in the timedomain, the amount of phase error change is converted into an amount ofthe residual symbol synchronization shift by using a filter performingsynchronization correction, and the received signal in the time domainis passed through the filter selected a filter coefficient correspondingto the amount of the symbol synchronization shift.
 23. The phase errorcorrection method according to claim 16, wherein in the time domain,after the correction of the residual carrier frequency offset isperformed, the correction of the residual symbol synchronization shiftis performed.
 24. The phase error correction method according to claim16, wherein in the time domain, after the correction of the residualsymbol synchronization shift is performed, the correction of theresidual carrier frequency offset is performed. 25-27. (canceled)